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1 lt13 0 2/lt13 0 2- 5 micropower high output current step-up adjustable and fixed 5v dc/dc converters figure 1. 2-cell to 5v/600ma dc/dc converter s f ea t u re d u escriptio n 5v at 600ma or 12v at 120ma from 2-cell supply n 200 m a quiescent current n logic controlled shutdown to 15 m a n low v cesat switch: 310mv at 2a typical n burst mode tm operation at light load n current mode operation for excellent line and load transient response n available in 8-lead so or pdip n operates with supply voltage as low as 2v the lt ? 1302/LT1302-5 are micropower step-up dc/dc converters that maintain high efficiency over a wide range of output current. they operate from a supply voltage as low as 2v and feature automatic shifting between burst mode operation at light load, and current mode operation at heavy load. the internal low loss npn power switch can handle current in excess of 2a and switch at frequencies up to 400khz. quiescent current is just 200 m a and can be further reduced to 15 m a in shutdown. available in 8-pin pdip or 8-pin so packaging, the lt1302/ LT1302-5 have the highest switch current rating of any similarly packaged switching regulators presently on the market. u s a o pp l ic at i , ltc and lt are registered trademarks of linear technology corporation. burst mode is a trademark of linear technology corporation. n notebook and palmtop computers n portable instruments n personal digital assistants n cellular telephones n flash memory u a o pp l ic at i ty p i ca l + nc c3 0.1 m f shutdown v in i t v c sw pgnd shdn sense gnd 2 r c 20k c c 0.01 m f c1 100 m f d1 + c2 100 m f l1 10 m h 2 cells lt1302 ?f01 c1 = c2 = sanyo os-con l1 = coiltronics ctx10-3 coilcraft do3316-103 output 5v 600ma 1 8 7 65 3 4 d1 = motorola mbrs130lt3 LT1302-5 load current (ma) 1 78 efficiency (%) 80 82 84 86 10 100 1000 lt1302 ?ta02 76 74 72 70 88 90 v in = 3v v in = 2.5v v in = 2v 2-cell to 5v converter efficiency
2 lt13 0 2/lt13 0 2- 5 a u g w a w u w a r b s o lu t exi t i s wu u package / o rder i for atio v in voltage ............................................................. 10v sw voltage ............................................................. 25v fb voltage .............................................................. 10v shdn voltage ......................................................... 10v v c voltage ................................................................ 4v i t voltage .................................................................. 4v maximum power dissipation ............................ 700mw operating temperature range .................... 0 c to 70 c storage temperature range ............... C 65 c to 150 c lead temperature (soldering, 10 sec)................. 300 c consult factory for industrial and military grade parts. order part number lt1302cn8 lt1302cs8 lt1302cn8-5 lt1302cs8-5 s8 part marking 1302 13025 symbol parameter conditions min typ max units i q quiescent current v shdn = 0.5v, v fb = 1.3v l 200 300 m a v shdn = 1.8v l 15 25 m a v in input voltage range 2.0 v l 2.2 8 v v fb feedback voltage (lt1302) v c = 0.4v l 1.22 1.24 1.26 v feedback pin bias current (lt1302) v fb = 1v 100 na output sense voltage (LT1302-5) v c = 0.4v l 4.85 5.05 5.25 v output ripple voltage (LT1302-5) v c = 0.4v 50 mv sense pin resistance to ground (LT1302-5) 420 k w v os offset voltage see block diagram 15 mv comparator hysteresis (note 1) 5 mv oscillator frequency current limit not asserted (note 2) 175 220 265 khz l 160 310 khz dc maximum duty cycle 75 86 95 % t on switch on time current limit not asserted 3.9 m s t off switch off time 0.7 m s output line regulation 2 < v in < 8v l 0.06 0.15 %/v v cesat switch saturation voltage i sw = 2a 310 400 mv l 475 mv switch leakage current v sw = 5v, switch off l 0.1 10 m a switch current limit v c = 0.4v (burst mode operation) 1 a v c = 1.25v (full power) (note 3) l 2.0 2.8 3.9 a error amplifier voltage gain 0.9v v c 1.2v, d v c / d v fb 50 75 v/ v v shdnh shutdown pin high l 1.8 v v shdnl shutdown pin low l 0.5 v i shdn shutdown pin bias current v shdn = 5v l 820 m a v shdn = 2v 3 m a v shdn = 0v l 0.1 1 m a i t pin resistance to ground 3.9 k w t a = 25 c, v in = 2.5v, unless otherwise noted. e lectr ic al c c hara terist ics c d the l denotes specifications which apply over the 0 c to 70 c temperature range. note 1: hysteresis is specified at dc. output ripple depends on capacitor size and esr. note 2: the lt1302 operates in a variable frequency mode. switching frequency depends on load inductance and operating conditions and may be above specified limits. note 3: minimum switch current 100% tested. maximum switch current guaranteed by design. 1 2 3 4 8 7 6 5 top view gnd v c shdn (sense*)fb pgnd sw v in i t n8 package 8-lead pdip s8 package 8-lead plastic so *fixed version pins 1 and 8 are internally connected in soic package t jmax = 125 c, q ja = 100 c/w (n8) t jmax = 125 c, q ja = 80 c/w (s8)
3 lt13 0 2/lt13 0 2- 5 typical perfor m a n ce characteristics uw temperature ( c) ?0 saturation voltage (mv) 400 350 300 250 200 150 100 ?5 02550 1302 g03 75 100 i sw = 2a temperature ( c) ?0 feedback voltage (v) 1.250 1.245 1.240 1.235 1.230 1.225 1.220 1.215 1.210 1.205 1.200 ?5 02550 1302 g04 75 100 supply voltage (v) 2.0 2.5 3.0 3.5 4.0 4.5 5.0 quiescent current ( m a) 500 450 400 350 300 250 200 150 100 50 0 1302 g01 t a = 25 c no-load quiescent current circuit of figure 1 temperature ( c) ?0 sense resistance (k w ) 600 500 400 300 200 100 0 ?5 02550 1302 g05 75 100 temperature ( c) ?0 quiescent current ( m a) 300 250 200 150 100 50 0 ?5 02550 1302 g06 75 100 v in = 2.5v switch off temperature ( c) ?0 offset voltage (mv) 30 25 20 15 10 5 0 ?5 02550 1302 g07 75 100 temperature ( c) ?0 output voltage (v) 5.100 5.075 5.050 5.025 5.000 4.975 4.950 4.925 4.900 ?5 02550 1302 g08 75 100 temperature ( c) ?0 on-time ( m s) 5.0 4.5 4.0 3.5 3.0 2.5 2.0 ?5 02550 1302 g09 75 100 switch current (a) 0 v cesat (v) 600 500 400 300 200 100 0 12 34 1302 g02 t a = 25 c switch saturation voltage switch saturation voltage lt1302 feedback voltage LT1302-5 sense pin resistance quiescent current error amplifier offset voltage LT1302-5 output voltage maximum on-time
4 lt13 0 2/lt13 0 2- 5 typical perfor m a n ce characteristics uw temperature ( c) ?0 duty cycle (%) 100 90 80 70 60 50 ?5 02550 1302 g10 75 100 temperature ( c) ?0 frequency (khz) 300 275 250 225 200 175 150 ?5 02550 1302 g11 75 100 shutdown voltage (v) 0 shutdown current ( m a) 20 18 16 14 12 10 8 6 4 2 0 4 5 1302 g12 13 2 6 7 8 t a = 25 c gnd (pin 1): signal ground. feedback resistor and 0.1 m f ceramic bypass capacitor from v in should be connected directly to this pin. v c (pin 2): frequency compensation pin. connect series rc to gnd. keep trace short. shdn (pin 3): shutdown. pull high to effect shutdown; tie to ground for normal operation. fb/sense (pin 4): feedback/sense. on the lt1302 this pin connects to cmp1 input. on the LT1302-5 this pin connects to the output resistor string. pi fu ctio s u uu i t (pin 5): normally left floating. addition of a 3.3k resistor to gnd forces the lt1302 into current mode at light loads. efficiency drops at light load but increases at medium loads. see applications information section. v in (pin 6): supply pin. must be bypassed with: (1) a 0.1 m f ceramic to gnd, and (2) a large value electrolytic to pgnd. when v in is greater than 5v, a low value resistor (2 w to 10 w ) is recommended to isolate the v in pin from input supply noise. maximum duty cycle oscillator frequency shutdown pin bias current LT1302-5 output voltage vs load current load current (a) 0 output voltage (v) 5.20 5.15 5.10 5.05 5.00 4.95 4.90 4.85 4.80 0.1 0.2 0.3 0.4 1302 g13 0.5 0.6 0.7 0.8 0.9 1.0 v in = 2.2v v in = 3v v in = 4v maximum output power* boost mode input voltage (v) 0 output power (w) 20 16 12 8 4 0 8 1302 g14 2 4 * approximate 6 10
5 lt13 0 2/lt13 0 2- 5 pi fu ctio s u uu sw (pin 7): switch pin. connect inductor and diode here. keep layout short and direct. pgnd (pin 8): power ground. pins 8 and 1 should be connected under the package. in the so package, pins 1 and 8 are thermally connected to the die. one square inch of pcb copper provides an adequate heat sink for the device. figure 2. lt1302 block diagram 1302 f02 + + + v os 15mv r2 c5 100pf 36mv cmp1 off v in 2 m a enable a2 v in c2 0.1 m f c1 v in l1 sw pgnd i t v c r3 22k c4 0.01 m f gnd shdn shutdown fb error amplifier hysteretic comparator driver a1 4 3 5 8 7 2 1 1.24v reference 220khz oscillator 300 w 3.6k q3 r4 1.75 w r5 730 w v in bias r1 6 + c3 d1 v out + q4 160x a3 q1 q2 q5 block diagra s m w
6 lt13 0 2/lt13 0 2- 5 block diagra s m w the lt1302s operation can best be understood by examining the block diagram in figure 2. the lt1302 operates in one of two modes, depending on load. with light loads, comparator cmp1 controls the output; with heavy loads, control is passed to error amplifier a1. burst mode operation consists of monitoring the fb pin voltage with hysteretic comparator cmp1. when the fb voltage, related to the output voltage by external attenu- ator r1 and r2, falls below the 1.24v reference voltage, the oscillator is enabled. switch q4 alternately turns on, causing current buildup in inductor l1, then turns off, allowing the built-up current to flow into output capaci- tor c3 via d1. as the output voltage increases, so does the fb voltage; when it exceeds the reference plus operatio u cmp1s hysteresis (about 5mv) cmp1 turns the oscilla- tor off. in this mode, peak switch current is limited to approximately 1a by a2, q2, and q3. q2s current, set at 34 m a, flows through r5, causing a2s negative input to be 25mv lower than v in . this node must fall more than 36mv below v in for a2 to trip and turn off the oscillator. the remaining 11mv is generated by q3s current flow- ing through r4. emitter-area scaling sets q3s collector current to 0.625% of switch q4s current. when q4s current is 1a, q3s current is 6.25ma, creating an 11mv drop across r4 which, added to r5s 25mv drop, is enough to trip a2. when the output load is increased to the point where the 1a peak current cannot support the output voltage, 1302 f03 + + + v os 15mv 36mv cmp1 off v in 2 m a enable a2 v in sw pgnd i t v c gnd shdn shutdown sense error amplifier hysteretic comparator driver a1 4 3 5 8 7 2 1 1.24v reference 220khz oscillator 300 w 3.6k q3 r4 1.75 w r5 730 w v in bias r1 315k r2 105k 6 q4 160x a3 q1 q2 q5 figure 3. LT1302-5 block diagram
7 lt13 0 2/lt13 0 2- 5 operatio u cmp1 stays on and the peak switch current is regulated by the voltage on the v c pin (a1s output). v c drives the base of q1. as the v c voltage rises, q2 conducts less current, resulting in less drop across r5. q4s peak current must then increase in order for a2 to trip. this current mode control results in good stability and immu- nity to input voltage variations. because this is a linear, closed-loop system, frequency compensation is required. a series rc from v c to ground provides the necessary pole-zero combination. the LT1302-5 incorporates feedback resistors r1 and r2 into the device. output voltage is set at 5.05v in burst mode, dropping to 4.97v in current mode. applicatio n s i n for m atio n wu u u inductor selection inductors used with the lt1302 must fulfill two require- ments. first, the inductor must be able to handle current of 2.5a to 3a without runaway saturation. rod or drum core units usually saturate gradually and it is acceptable to exceed manufacturers published saturation currents by 20% or so. second, it should have low dcr, under 0.05 w so that copper loss is kept low. inductance value is not critical. generally, for low voltage inputs down to 2v, a 10 m h inductor is recommended (such as coilcraft do3316- 103). for inputs above 4v to 5v use a 22 m h unit (such as coilcraft do3316-223). switching frequency can reach up to 400khz so the core material should be able to handle high frequency without loss. ferrite or molypermalloy cores are a better choice than powdered iron. if emi is a concern a toroidal inductor is suggested, such as coiltronics ctx20-4. for a boost converter, duty cycle can be calculated by the following formula: dc = 1 v v in out ? ? ? ? a special situation exists where the v out /v in differential is high, such as a 2v-to-12v converter. the required duty cycle is higher than the lt1302 can provide, so the converter must be designed for discontinuous operation. this means that inductor current goes to zero during the switch off-time. in the 2v-to-12v case, inductance must be low enough so that current in the inductor can reach 2a in a single cycle. inductor value can be defined by: l vv t a in sw on - () 2 with the 2v input a value of 3.3 m h is acceptable. since the inductance is so low, usually a smaller core size can be used. efficiency will not be as high as for the continuous case since peak currents will necessarily be higher. table 1 lists inductor suppliers along with appropriate part numbers. table 1. recommended inductors vendor part no. value( m h) phone no. coilcraft do3316-103 10 (708) 639-6400 do3316-153 15 do3316-223 22 coiltronics ctx10-2 10 (407) 241-7876 ctx20-4 20 dale lpt4545-100la 10 (605) 665-9301 lpt4545-200la 20 sumida cd105-100 10 (708) 956-0666 cd105-150 15 cdr125-220 22 capacitor selection the output capacitor should have low esr for proper performance. a high esr capacitor can result in mode- hopping between current mode and burst mode at high load currents because the output voltage will increase by i sw esr when the inductor current is flowing into the diode. figure 4 shows output voltage of an LT1302-5 boost converter with two 220 m f avx tps capacitors at the output. ripple voltage at a 510ma load is about 30mv p-p
8 lt13 0 2/lt13 0 2- 5 applicatio n s i n for m atio n wu u u and there is no low frequency component. the total esr is under 0.03 w . if a single 100 m f aluminum electrolytic capacitor is used instead, the converter mode-hops be- tween current mode and burst mode due to high esr, causing the voltage comparator to trip as shown in figure 5. the ripple voltage is now over 500mv p-p and contains a low frequency component. maximum allowable output capacitor esr can be calculated by the following formula: esr vv va max os out ref = 1 where, v os = 15mv v ref = 1.24v 500 m s/div 1302 f04 v out 50mv/div ac coupled i load figure 4. low esr output capacitor results in stable operation. ripple voltage is under 30mv p-p 500 m s/div 1302 f05 v out 200mv/div ac coupled i load figure 5. inexpensive electrolytic capacitor has high esr, resulting in mode-hop, ripple voltage amplitude is over 500mv p-p and includes low frequency component 510ma 10ma 510ma 10ma input capacitor the input supply should be decoupled with a good quality electrolytic capacitor close to the lt1302 to provide a stable input supply. long leads or traces from power source to the switcher can have considerable impedance at the lt1302s switching frequency. the input capacitor provides a low impedance at high frequency. a 0.1 m f ceramic capacitor is required right at the v in pin. when the input voltage can be above 5v, a 10 w /1 m f decoupling network for v in is recommended as detailed in figure 6. this network is also recommended when driving a trans- former. figure 6. a 10 w /1 m f decoupling network at v in is recommended when input voltage is above 5v + + v in sw pgnd 1 m f lt1302 ? ?? gnd 47 m f to 100 m f 10 w 1302 f06 v in > 5v table 2 lists capacitor vendors along with device types. table 2. recommended capacitors vendor series type phone no. avx tps surface mount (803) 448-9411 sanyo os-con through hole (619) 661-6835 sprague 595d surface mount (603) 224-1961 diode selection a 2a schottky diode such as motorola mbrs130lt3 has been found to be the best available. other choices include 1n5821 or mbrs130t3. do not use general purpose diodes such as 1n4001. they are much too slow for use in switching regulator applications.
9 lt13 0 2/lt13 0 2- 5 applicatio n s i n for m atio n wu u u behavior in the 4th graticule is the result of the lt1302s burst mode comparator turning off all switching as output voltage rises above its threshold. in figure 7c, the 0.1 m f capacitor has been replaced by a 0.01 m f unit. undershoot is less but the response is still underdamped. figure 7d shows the results of the 0.1 m f capacitor and a 10k resistor in series. now some amount of damping is observed, and behavior is more controlled. figure 7e details response with a 0.01 m f/10k series net- work. undershoot is down to around 100mv, or 2%. a slight underdamping is still noticeable. finally, a 0.01 m f/24k series network results in the re- sponse shown in figure 7f. this has optimal damping, undershoot less than 100mv and settles in less than 1ms. the v c pin is sensitive to high frequency noise. some layouts may inject enough noise to modulate peak switch current at 1/2 the switching frequency. a small capacitor connected from v c to ground will eliminate this. do not exceed 1/10 of the compensation capacitor value. frequency compensation obtaining proper rc values for the frequency compensa- tion network is largely an empirical procedure, since variations in input and output voltage, topology, capacitor esr and inductance make a simple formula elusive. as an example, consider the case of a 2.5v to 5v boost converter supplying 500ma. to determine optimum compensation, the circuit is built and a transient load is applied to the circuit. figure 7 shows the setup. in figure 7a, the v c pin is simply left floating. although output voltage is maintained and transient response is good, switch current rises instantaneously to the internal current limit upon application of load. this is an undesir- able situation as it places maximum stress on the switch and the other power components. additionally, efficiency is well down from its optimal value. next, a 0.1 m f capacitor is connected with no resistor. figure 7b details response. although the circuit eventually stabilizes, the loop is quite underdamped. initial output sag exceeds 5%. aberrant figure 7a. v c pin left unconnected. output shows low frequency components under load figure 7b. 0.1 m f from v c to ground. better, but more improvement needed 2ms/div 1302 f07b v out 100mv/div ac coupled i load 510ma 10ma v out 100mv/div ac coupled 2ms/div 1302 f07a i load 510ma 10ma figure 7. boost converter with simulated load v in i t shdn sense sw l1 10 m h d1 v c mtp3055el r c pulse generator pgnd 0.1 m f LT1302-5 gnd 10 w 2w 500 w nc 1302 f07 v in 2.5v 50 w + c3 220 m f + c2 220 m f c1, c2, c3 = avx tps series d1 = motorola mbrs130lt3 l1 = coilcraft do3316-103k + c1 330 m f
10 lt13 0 2/lt13 0 2- 5 applicatio n s i n for m atio n wu u u 2ms/div 1302 f07c v out 100mv/div ac coupled i load figure 7c. 0.01 m f from v c to ground. underdamped response requires series r 510ma 10ma i t pin the i t pin is used to disable burst mode, forcing the lt1302 to operate in current mode even at light load. to disable burst mode, 3.3k resistor r1 is connected from i t to gound. more conservative frequency compensation must be used when in this mode. a 0.1 m f capacitor and 4.7k resistor from v c to ground has been found to be adequate. low frequency burst mode ripple can be reduced or eliminated using this technique in many appli- cations. to illustrate, the transient load response of figure 8s circuit is pictured without and with r1. figure 8a shows output voltage and inductor current without the resistor. note the 6khz burst rate when the converter is delivering 25ma. by adding the 3.3k resistor, the low frequency bursting is eliminated, as shown in figure 8b. this feature is useful in systems that contain audio circuitry. at very light or zero load, switching frequency drops and eventu- 1ms/div 1302 f08a v out 100mv/div ac coupled figure 8a. i t pin floating. note 6khz burst rate at i load = 25ma. 0.1 m f/4.7k compensation network causes 220mv undershoot 525ma 25ma i load inductor current 1a/div v out 100mv/div ac coupled 2ms/div 1302 f07f 510ma 10ma i load figure 7f. 0.01 m f, 24k series rc results in optimum response 2ms/div 1302 f07e v out 100mv/div ac coupled i load 510ma 10ma figure 7e. 0.01 m f, 10k series rc shows good transient response. slight underdamping still noticeable 2ms/div 1302 f07d v out 100mv/div ac coupled i load figure 7d. 0.1 m f with 10k series rc. classic overdamped response 510ma 10ma v in sense v c sw 10 m h v out 5v 600ma i t mbrs130lt3 pgnd 0.1 m f LT1302-5 gnd 4.7k r1 3.3k 1302 f08 v in 2.5v + 220 m f 10v + 220 m f 10v + c1 330 m f 0.1 m f figure 8. addition of r1 eliminates low frequency output ripple in this 2.5v to 5v boost converter
11 lt13 0 2/lt13 0 2- 5 applicatio n s i n for m atio n wu u u 1ms/div 1302 f08b v out 100mv/div ac coupled figure 8b. 3.3k resistor from i t pin to ground forces lt1302 into current mode regardless of load. audio frequency component eliminated 525ma 25ma i load inductor current 1a/div the i t pin cannot be used as a soft-start. large capacitors connected to the pin will cause erratic operation. if oper- ating the device in burst mode, let the pin float. keep high dv/dt signals away from the pin. figure 8c details efficiency with and without the addition of r1. burst mode operation keeps efficiency high at light load with i t floating. efficiency falls off at light load with r1 added because the lt1302 cannot transition into burst mode. layout the high speed, high current switching associated with the lt1302 mandates careful attention to layout. follow the suggested component placement in figure 9 for proper operation. high current functions are separated by the package from sensitive control functions. feedback resis- tors r1 and r2 should be close to the feedback pin (pin4). noise can easily be coupled into this pin if care is not taken. a small capacitor (100pf to 200pf) from fb to ground provides a high frequency bypass. if the lt1302 is oper- ated off a three-cell or higher input, r3 (2 w to 10 w ) in series with v in is recommended. this isolates the device from noise spikes on the input supply. do not put in r3 if the device must operate from a 2v input, as input current will cause the voltage at the lt1302s v in pin to go below 2v. the 0.1 m f ceramic bypass capacitor c3 (use x7r, not z5u) should be mounted as close as possible to the package. when r3 is used, c3 should be a 1 m f tantalum unit. grounding should be segregated as illustrated. c3s ground trace should not carry switch current. run a figure 8c. 3.3k resistor for i t to ground increases efficiency at moderate load, decreases at light load output current (ma) 1 efficiency (%) 90 80 70 60 50 40 30 10 100 1000 1302 f08c i t floating 3.3k i t to gnd figure 9. suggested component placement for lt1302 ally reaches audio frequencies, but at a much lighter load than without the i t feature. at some input voltage/load current combinations, some residual bursting may occur at frequencies out of the audio band. 1302 f09 4 3 2 1 8 7 6 5 lt1302 c3 r3 2 w d1 c2 c1 v out v in l1 + + 200pf r1 r2 r c c c shutdown gnd (battery and load return)
12 lt13 0 2/lt13 0 2- 5 applicatio n s i n for m atio n wu u u table 3. s8 package, 8-lead plastic so copper area thermal resistance topside* backside board area (junction-to-ambient) 2500 sq. mm 2500 sq. mm 2500 sq. mm 60 c/w 1000 sq. mm 2500 sq. mm 2500 sq. mm 62 c/w 225 sq. mm 2500 sq. mm 2500 sq. mm 65 c/w 100 sq. mm 2500 sq. mm 2500 sq. mm 69 c/w 100 sq. mm 1000 sq. mm 2500 sq. mm 73 c/w 100 sq. mm 225 sq. mm 2500 sq. mm 80 c/w 100 sq. mm 100 sq. mm 2500 sq. mm 83 c/w * pins 1 and 8 attached to topside copper n8 package, 8-lead dip: thermal resistance (junction-to-ambient) = 100 c/w calculating temperature rise power dissipation internal to the lt1302 in a boost regulator configuration is approximately equal to: the first term in this equation is due to switch on- resistance. the second term is from the switch driver. r is switch resistance, typically 0.15 w . v d is the diode forward drop. the temperature rise can be calculated from: d t = p d q ja where: d t = temperature rise p d = device power dissipation q ja = thermal resistance (junction-to-ambient) pir vv v ivr v vv v ivr v iv vv d out out d in out out in out d in out out in out out d in = + - ? ? ? ? ? ? ? - + - ? ? ? ? ? ? ? ? ? + +- () 2 2 27 separate ground trace up under the package as shown. the battery and load return should go to the power side of the ground copper. thermal considerations the lt1302 contains a thermal shutdown feature which protects against excessive internal (junction) tempera- ture. if the junction temperature of the device exceeds the protection threshold, the device will begin cycling be- tween normal operation and an off state. the cycling is not harmful to the part. the thermal cycling occurs at a slow rate, typically 10ms to several seconds, which depends on the power dissipation and the thermal time constants of the package and heat sinking. raising the ambient tem- perature until the device begins thermal shutdown gives a good indication of how much margin there is in the thermal design. for surface mount devices heat sinking is accomplished by using the heat spreading capabilities of the pc board and its copper traces. experiments have shown that the heat spreading copper layer does not need to be electri- cally connected to the tab of the device. the pcb material can be very effective at transmitting heat between the pad area attached to pins 1 and 8 of the device, and a ground or power plane layer either inside or on the opposite side of the board. although the actual thermal resistance of the pcb material is high, the length/area ratio of the thermal resistance between the layer is small. copper board stiff- eners and plated through holes can also be used to spread the heat generated by the device. table 3 lists thermal resistance for the so package. measured values of thermal resistance for several differ- ent board sizes and copper areas are listed for each surface mount package. all measurements were taken in still air on 3/32 " fr-4 board with 1oz copper. this data can be used as a rough guideline in estimating thermal resis- tance. the thermal resistance for each application will be affected by thermal interactions with other components as well as board size and shape.
13 lt13 0 2/lt13 0 2- 5 applicatio n s i n for m atio n wu u u as an example, consider a boost converter with the following specifications: v in = 3v v out = 6v i out = 700ma total power loss in the lt1302, assuming r = 0.15 w and v d = 0.45v, is: using the cs8 package with 100 sq. mm topside and backside heat sinking: d t = (312mw)(84 c/w) = 25.9 c rise with the n8 package: d t = 31.2 c at a 70 c ambient, die temperature would be 101.2 c. pma mw mw mw d = ()() + - ? ? ? ? ? ? ? - + - ? ? ? ? ? ? ? ? ? + () +- () =+= 700 0 15 6045 3 07 6 015 3 6045 3 07 6 015 3 0 7 6 0 45 3 27 223 89 312 2 2 . . .. . .. .. w
14 lt13 0 2/lt13 0 2- 5 typical applicatio n s u single cell to 5v/150ma converter 1302 ta03 1.5v cell 100k 100k 0.01 m f 20k 100pf 0.1 m f c2 220 m f c1 47 m f l1 = coilcraft do3316-332 d1 = motorola mbrs130lt3 c1 = avx tpsd476m016r0150 c2 = avx tpse227m010r0100 coilcraft (708) 639-2361 100k 56.2k 1% 4.99k 1% 36.5k 1% r1 301k 1% (169k for 3.3v) 10 w d1 5v/150ma output l1 3.3 m h 220 w + + lt1302 fb gnd pgnd shdn v in sw i t v c sw1 a o sw2 gnd v in i l set fb lt1073 2n3906 2v to 12v/120ma converter + v in i t nc v c sw c3 0.1 m f lt1302 pgnd shdn shutdown fb gnd 2 r c 20k r1 100k 1% r2 866k 1% c c 0.02 m f c1 100 m f d1 + c2 33 m f c2 33 m f l1 3.3 m h 2 cells lt1302 ?ta04 c1 = avx tpsd107m010r0100 c2 = avx tpsd336m025r0200 output 12v 120ma 100pf 1 8 7 65 3 4 d1 = motorola mbrs130lt3 l1 = coilcraft do3316-332 +
15 lt13 0 2/lt13 0 2- 5 information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. typical applicatio n s u 2 li-ion cell to 5.8v/600ma dc/dc converter 3 cell to 3.3v buck-boost converter with auxiliary 12v regulated output 1302 ta05 lt1302 gnd pgnd shdn v in v c fb i t sw shutdown 169k 1% 200pf 100k 1% 24k 4700pf 10 w 0.1 m f 3 8 t1c 1 10 t1a 7 4 t1d 6 5 t1e v in 2.5v-8v c3 47 m f 16v + 2 9 t1b 22 m f 25v + c1 100 m f 16v + d1 d2 13v c2 330 m f 6.3v + 3.3v output 400ma 330k 1% 3.3 m f + 150k 1% 12v 120ma t1 = d1, d2 = c1 = c2 = c3 = dale lpe-6562-a069, 1:3:1:1:1 turns ratio, 10 m h primary. dale (605) 665-9301 motorola mbrs130lt3 avx tpse107016r0100 avx tpse337006r0100 avx tpsd476016r0150 lt1121 gnd in shdn adj out v in i t fb shdn shutdown l1 22 m h l2 22 m h v c pgnd 1 m f 10 w lt1302 gnd sw 20k 1302 ta07 v out 5.8v 600ma v in 4v to 9v 100k 1% mbrs130lt3 365k 1% + ++ c1 100 m f 16v + c2 220 m f 10v c3 220 m f 10v 10nf duty cycle = v out v in + v out peak switch voltage = v in + v out l1, l2 = coilcraft do3316-223 c1 = avx tpse107016r0100 c2, c3 = avx tpse227010r0100
16 lt13 0 2/lt13 0 2- 5 linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7487 (408) 432-1900 l fax : (408) 434-0507 l telex : 499-3977 ? linear technology corporation 1995 lt/gp 0295 10k ? printed in usa package descriptio n u dimensions in inches (millimeters) unless otherwise noted. n8 0694 0.045 0.015 (1.143 0.381) 0.100 0.010 (2.540 0.254) 0.065 (1.651) typ 0.045 ?0.065 (1.143 ?1.651) 0.130 0.005 (3.302 0.127) 0.015 (0.380) min 0.018 0.003 (0.457 0.076) 0.125 (3.175) min 12 3 4 87 6 5 0.255 0.015* (6.477 0.381) 0.400* (10.160) max 0.009 ?0.015 (0.229 ?0.381) 0.300 ?0.325 (7.620 ?8.255) 0.325 +0.025 0.015 +0.635 0.381 8.255 () *these dimensions do not include mold flash or protrusions. mold flash or protursions shall not exceed 0.010 inch (0.254mm). 1 2 3 4 0.150 ?0.157* (3.810 ?3.988) 8 7 6 5 0.189 ?0.197* (4.801 ?5.004) 0.228 ?0.244 (5.791 ?6.197) 0.016 ?0.050 0.406 ?1.270 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) so8 0294 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) bsc *these dimensions do not include mold flash or protrusions. mold flash or protrusions shall not exceed 0.006 inch (0.15mm). n8 package 8-lead plastic dip s8 package 8-lead plastic soic

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