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MIC3223
High Power Boost LED Driver with Integrated FET
General Description
The MIC3223 is a constant current boost LED driver capable of driving a series string of high power LEDs. The MIC3223 can be used in general lighting, bulb replacement, garden pathway lighting and other solid state illumination applications. The MIC3223 is a peak current mode control PWM boost regulator and the 4.5V and 20V operating input voltage range allows multiple applications from a 5V or a 12V bus. The MIC3223 implements a fixed internal 1MHz switching frequency to allow for a reduction in the design footprint size. Power consumption has been minimized through the implementation of a 200mV feedback voltage that provides an accuracy of 5%. The MIC3223 can be dimmed through the use of a PWM signal and features an enable pin for a low power shutdown state. The MIC3223 is a very robust LED driver and offers the following protection features: over voltage protection (OVP), thermal shutdown, switch current limiting and under voltage lockout (UVLO). The MIC3223 is offered in a low profile exposed pad 16-pin TSSOP package. Data sheets and support documentation can be found on Micrel's web site at: www.micrel.com.
Features
* * * * * * * * * 4.5V to 20V supply voltage 200mV feedback voltage with an accuracy of 5% Step-up output voltage (boost) conversion up to 37V 1MHz switching frequency 100m/3.5A internal power FET switch LEDs can be dimmed using a PWM signal User settable LED current (through external resistor) Externally programmable soft-start Protection features that include: - Output over-voltage protection (OVP) - Under-voltage lockout (UVLO) - Over temperature protection * Junction temperature range: -40C to +125C * Available in a exposed pad 16-pin TSSOP package
Applications
* * * * * * Architectural lighting Industrial lighting Signage Landscape lighting (garden/pathway) Under cabinet lighting MR-16 bulbs
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Typical Application
Micrel Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel +1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com
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Ordering Information
Part Number MIC3223YTSE Junction Temp. Range -40 to +125C Package 16-pin ePad TSSOP Lead Finish PB- free
Pin Configuration
16-Pin ePad TSSOP (TSE)
Pin Description
Pin Number 1 2 Pin Name EN SS Pin Function Enable (Input): Logic high enables and logic low disables operation. Soft Start (Input resistance of 30k). Connect a capacitor to GND for soft-start. Clamp the pin to a known voltage to control the internal reference voltage and hence the output current. Compensation Pin (Input): Add external R and C-to-GND to stabilize the converter. Negative Input to Error Amp Connect to the centre tap of an external resistor divider, the top of which is tied to Vout and bottom-to-ground. Power Ground Switch Node (Input): Internal NMOS switch Drain Pin Input Supply For 4.5V < VIN < 6V, connect DRVVDD to VIN. DRVVDD is the input voltage supply for the converter's internal power FET gate driver. For VIN > 6V, connect this pin to VDD. For 4.5V < VIN < 6V, this pin becomes the input voltage supply for the converter's internal circuit. For VIN > 6V, this pin is an output of the internal 5.5V regulator that supplies internal circuits. User must add 10F decoupling capacitor from VDD-to-AGND. PWM input to control LED dimming. Output driver to drive external FET for LED dimming. Analog Ground Connect to Power Ground
3 4 5 6 7,8,9,10 11 12 13
COMP FB OVP PGND SW VIN DRVVDD VDD
14 15 16 17
DIM_IN DIM_OUT AGND EP
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Absolute Maximum Ratings(1)
Supply Voltage (VIN) .....................................................+22V Switch Voltage (VSW)..................................... -0.3V to +42V Regulated Voltage (VDD) ............................... -0.3V to +6.5V Dimming In Voltage (VDIM_IN) ...............-0.3V to (VDD + 0.3V) Dimming Out Voltage (VDIM_OUT)..........-0.3V to (VDD + 0.3V) Soft-Start Voltage (VSS) .......................-0.3V to (VDD + 0.3V) Enable Voltage (VEN)............................-0.3V to (VIN + 0.3V) Feedback Voltage (VFB) ......................-0.3V to (VDD + 0.3V) Switch Current (ISW) ..................................Internally Limited Comp Voltage (VCOMP).......................-0.3V to (+VDD + 0.3V) FET Driver Supply (VDRVVDD) ......................... -0.3V to +6.5V PGND to AGND ............................................ -0.3V to +0.3V Over Voltage Protection (VOVP) ...........-0.3V to (VDD + 0.3V) Peak Reflow Temperature (soldering, 10-20sec.) ..... 260C Storage Temperature (TS)..........................-65C to +150C ESD Rating(3) ................................................................+2kV
Operating Ratings(2)
Supply Voltage (VIN)...................................... +4.5V to +20V Switch Voltage (VSW)....................................................+37V Junction Temperature (TJ) .........................-40C to +125C Junction Thermal Resistance ePad TSSOP-16L (JA)...................................36.5C/W
Electrical Characteristics(4)
VIN = VEN = 12V; L = 22H, CIN =4.7F, COUT =2x4.7F; TA = 25C, BOLD values indicate -40C TJ +125C, unless otherwise noted. Symbol VIN VUVLO VOVP IVIN ISD VFB IFB VDD DMAX ISW RSW ISW VEN IEN VDIM_TH_H VDIM_TH_L Hys IDIM_IN TDR TDF Parameter Voltage Supply Range Under Voltage Lockout Over Voltage Protection Quiescent Current Shutdown Current Feedback Voltage Feedback Input Current Internal Voltage Regulator Maximum Duty Cycle VDD Line Regulation Switch Current Limit Switch RDSON plus RCS Switch Leakage Current Enable Threshold Enable Pin Current DIM_IN Threshold High DIM_IN Threshold Low DIM_IN Hysteresis DIM_IN Pin Current Dim Delay (Rising) Dim Delay (Falling VDIM_IN = 5V DIM_IN Rising DIM_IN Falling 40 30 Logic High Logic Low 500 1 1.5 0.4 VEN=0, VSW=37V Turn On Turn Off 20 1.5 0.4 40 VLED=18V, VIN=8V to 16V, ILED=350mA 3.5 85 VFB=250mV VEN =0V Room Temperature Over Temperature VFB=200mV 190 184 -450 5.3 90 0.5 9 100 0.01 10 10.5 95 200 Monitoring for VDD Condition Min 4.5 3 1.216 3.7 1.28 2.1 Typ Max 20 4.4 1.344 5 10 210 216 Units V V V mA A mV mV nA V % % A m A V V A V V mV A ns ns
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Symbol DIM MIN Parameter Minimum Dimming Pulse Condition DIM_IN =1s CDIM_OUT = 1.25nF DIM_OUT measured from 4V rising to 2.5 falling DIM_OUT pull up resistance IDIM_OUT = +2mA Dim Out pull down resistance IDIM_OUT = -2mA 0.7 30 Temperature rising Hysteresis Min 0.7 0.5 70 40 1 46 165 10 1.3 62 Typ Max 1.3 1.5
MIC3223
Units s s MHz k C C
RDO RDO FSW RSS TSD
Notes
DIM_OUT Resistance High DIM_OUT Resistance Low Oscillator Frequency Soft Start Resistance Over Temperature Threshold Shutdown
1. Exceeding the absolute maximum rating may damage the device. 2. The device is not guaranteed to function outside its operating rating. 3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF. 4. Specification for packaged product only.
Test Circuit
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Typical Characteristics
Efficiency v s. Input Voltage
98 96 94 EFFICIENCY (%) 92 90 88 86 84 82 80 5 10 15 20 INPUT VOLT AGE (V) VOUT = 25V IOUT = 0.5A T = 25C
VDD Voltage v s. Input Voltage
5.50 5.45 5.40 VDD VOLTAGE (V) 5.35 5.30 5.25 5.20 5.15 5.10 5.05 5.00 5 10 15 20 INPUT VOLT AGE (V) VOUT = 25V IOUT = 0.5A T = 25C CURRENT LIM IT (A) 9.0 9.5
Current Lim it v s. Input Voltage
T = 25C 8.5 8.0 7.5 VIN = 4.5V to 6V 7.0 4 9 14 19 INPUT VOLT AGE (V)
Feedback Voltage v s. Input Voltage
0.210 0.208
SWITCHING FREQUENCY (MHz)
Switching Frequency v s. Input Voltage
1.2 T = 25C
Feedback Voltage v s. Temperature
0.220 0.218 FEEDBACK VOLTAGE (V) 0.216 0.214 0.212 0.210 0.208 0.206 0.204 0.202 0.200 -40 -20 0 20 40 60 VIN = 12V VOUT = 26V IOUT = 0.36A 80 100 120
REFERENCE VOTLAG E (V)
0.206 0.204 0.202 0.200 0.198 0.196 0.194 0.192 0.190 4 9 14 19 INPUT VOLT AG E (V) VOUT = 30V IOUT =0.36A
1.1
1.1 1.0
VDD = VIN VIN = 4.5V to 6V VOUT = 30V IOUT = 0.36A
1.0
0.9 4 9 14 19 INPUT VOLT AGE (V)
T EM PERAT URE (C)
Current Lim it v s. Temperature
11.0 10.5 10.0 CURRENT LIM IT (A)
RSW_NO DE () 0.18 0.17 0.16 0.15 0.14 0.13 0.12 0.11 0.10
RSW _NODE vs. T emperature
1.20 1.15
SWITCHING FREQUENCY (MHz)
Switching Frequency v s. Temperature
9.5 9.0 8.5 8.0 7.5 7.0 6.5 6.0 -40 -20 0 20 40 60 80 100 120 T EM PERAT URE (C) VIN = 12V
1.10 1.05 1.00 0.95 0.90 0.85 0.80 -40 -20 0 20 40 60 80 100 120 T EM PERAT URE (C) VIN = 12V VOUT = 26V IOUT = 0.36A
VIN = 12V VOUT = 36V ISW = 1.3A -40 -20 0 20 40 60 80 100 120
T EM PERAT URE (C)
Efficiency v s. Output Current
96 94 92 EFFICIENCY (%) 90 88 86 84 82 80 0 0.5 1 1.5 OUT PUT CURRENT (A) VOUT = 25V 10V
EFFICIENCY (%) 96
Efficiency v s. Output Current
98
16V 14V 94 92
Efficiency v s. Output Current
96 94 EFFICIENCY (%) 92 90 88 86 84 18V 20V
12V
8V
90 88 86 84 82 80 0 0.5 1 1.5 OUT PUT CURRENT (A) VOUT = 25V
82 80 0 0.5 1
VOUT = 25V 1.5
OUT PUT CURRENT (A)
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Typical Characteristics (continued)
Efficiency v s. Output Current
96 94 92 EFFICIENCY (%) 90 88 86 84 82 80 0 0.5 1 1.5 OUT PUT CURRENT (A) VOUT = 25V EFFICIENCY (%) 10V 96 94 92 90 88 86 84 82 80 0 0.5 1 1.5 OUT PUT CURRENT (A) VOUT = 25V 12V
Efficiency v s. Output Current
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Functional Characteristics
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Functional Characteristics (continued)
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Functional Diagram
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MIC3223 current is regulated. If VFB drops, VEA increases and therefore the power FET remains on longer so that VCS can increase to the level of VEA. The reverse occurs when VFB increases. PWM Dimming This control process just described occurs during each DIM_IN pulse and when ever DIM_IN is high. When DIM_IN is low, the boost converter will no longer switch and the output voltage will drop. For high dimming ratios use an external PWM Dimming switch as shown in the Typical Application. When the dim pulse is on the external switch is on and circuit operates in the closed loop control mode as described. When the DIM_IN is low the boost converter does not switch and the external switch is open and no LED current can flow and the output voltage does not droop. When DIM_IN goes high the external switch is driven on and LED current flows. The output voltage remains the same (about the same) during each on and off DIM_IN pulse. PWM Dimming can also be used in the Test Circuit in applications that do not require high dimming ratios. In the Test Circuit, the load is not removed from the output voltage between DIM_IN pulses and will therefore drain the output capacitors. The voltage that the output will discharge to is determined by the sum of the VF (forward voltage drops of the LEDs). When VOUT can no longer forward bias the LEDs, then the LED current will stop and the output capacitors will stop discharging. During the next DIM_IN pulse VOUT has to charge back up before the full LED current will flow. For applications that do not require high dimming ratios.
Functional Description
A constant current output converter is the preferred method for driving LEDs. Small variations in current have a minimal effect on the light output, whereas small variations in voltage have a significant impact on light output. The MIC3223 LED driver is specifically designed to operate as a constant current LED Driver. The MIC3223 is designed to operate as a boost converter, where the output voltage is greater than the input voltage. This configuration allows for the design of driving multiple LEDs in series to help maintain color and brightness. The MIC3223 can also be configured as a SEPIC converter, where the output voltage can be either above or below the input voltage. The MIC3223 has an input voltage range, from 4.5V and 20V, to address a diverse range of applications. In addition, the LED current can be programmed to a wide range of values through the use of an external resistor. This provides design flexibility in adjusting the current for a particular application need. The MIC3223 features a low impedance gate driver capable of switching large MOSFETs. This low impedance provides higher operating efficiency. The MIC3223 can control the brightness of the LEDs via its PWM dimming capability. Applying a PWM signal (up to 20kHz) to the DIM_IN pin allows for control of the brightness of the LEDs. The MIC3223 boost converter employs peak current mode control. Peak current mode control offers advantages over voltage mode control in the following manner. Current mode control can achieve a superior line transient performance compared to voltage mode control and is easier to compensate than voltage mode control, thus allowing for a less complex control loop stability design. Page 9 of this datasheet shows the functional block diagram. Boost Converter operation The boost converter is a peak current mode pulse width modulation (PWM) converter and operates as follows. A flip-flop (FF) is set on the leading edge of the clock cycle. When the FF is set, a gate driver drives the power FET on. Current flows from VIN through the inductor (L) and through the power switch and also through the current sense resistor to PGND. The voltage across the current sense resistor is added to a slope compensation ramp (needed for stability). The sum of the current sense voltage and the slope compensation voltages (called VCS) is fed into the positive terminal of the PWM comparator. The other input to the PWM comparator is the error amp output (called VEA). The error amp's negative input is the feedback voltage (VFB). VFB is the voltage across RADJ (R5). In this way the output LED January 2010 10
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MIC3223 Output Over Voltage Protection (OVP) The MIC3223 provides an OVP circuitry in order to protect the system from an overvoltage fault condition. This OVP threshold can be programmed through the use of external resistors (R3 and R4 in the Typical Application). A reference value of 1.245V is used for the OVP. Equation 3 can be used to calculate the resistor value for R9 to set the OVP point. Normally use 100k for R3. Eq. (3)
R4 = R3 (VOVP /1.245) - 1
Application Information
Constant Output Current Converter The MIC3223 is a peak current mode boost converter designed to drive high power LEDs with a constant current output. The MIC3223 operates with an input voltage range from 4.5V to 20V. In the boost configuration, the output can be set from VIN up to 37V. The peak current mode control architecture of the MIC3223 provides the advantages of superior line transient response as well as an easier to design compensation. The MIC3223 LED driver features a built-in soft start circuitry in order to prevent start-up surges. Other protection features include: * Current Limit (ILIMIT) - Current sensing for over current and overload protection * Over Voltage Protection (OVP) - output over voltage protection to prevent operation above a safe upper limit * Under Voltage Lockout (UVLO) - UVLO designed to prevent operation below a safe lower limit Setting the LED Current The current through the LED string is set via the value chosen for the current sense resistor RADJ which is R5 in the schematic of the Typical Application. This value can be calculated using Equation 1: Eq. (1)
ILED = 0.2V R ADJ
VDD An internal linear regulator is used to provide the necessary internal bias voltages. When VIN is 6V or below connect the VDD pin to VIN. Use a 10F ceramic bypass capacitor. DRVVDD An internal linear regulator is used to provide the necessary internal bias voltages to the gate driver that drives the external FET. When VIN is above 6V connect DRVVDD to VDD. When VIN is 6V or below connect the DRVVDD pin to VIN. Use a bypass capacitor, 10F ceramic capacitor. UVLO Internal under voltage lock out (UVLO) prevents the part from being used below a safe VIN voltage. The UVLO is 3.7V. Operation below 4.5V is not recommended. Soft Start Soft start is employed to lessen the inrush currents during turn on. At turn on the following occurs; 1. After about 1.5ms CSS will start to rise in a exponential manner according to;
VSS
-t (37kxC SS ) = 0.21 - e
Another important parameter to be aware of in the boost converter design is the ripple current. The amount of ripple current through the LED string is equal to the output ripple voltage divided by the LED AC resistance (RLED - provided by the LED manufacturer) plus the current sense resistor RADJ. The amount of allowable ripple through the LED string is dependent upon the application and is left to the designer's discretion. The equation is shown in Equation 2. Eq. (2) Where
ILED VOUTRIPPLE (RLED + R ADJ ) ILED x D COUT x FSW
VOUTRIPPLE =
Reference Voltage The voltage feedback loop the MIC3223 uses an internal voltage of 200mV with an accuracy of 5%. The feedback voltage is the voltage drop across the current sense resistor as shown in the Typical Application. When in regulation the voltage at VFB will equal 200mV. January 2010 11
2. According to the block diagram, VSS is the ref node of the error amp. PWM switching start when VSS begins to rise. 3. When the CSS is fully charged, 0.2V will be at the error amp reference and steady state operation begins. 4. Design for soft-start time using the above equation.
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MIC3223 If high dimming ratios are required, a lower Dimming frequency is required. During each DIM_IN pulse the inductor current has to ramp up to it steady state value in order for the programmed LED current to flow. The smaller the inductance value the faster this time is and a narrower DIM_IN pulse can be achieved. But smaller inductance means higher ripple current.
Figure 1. Soft start
LED Dimming The MIC3223 LED driver can control the brightness of the LED string via the use of pulse width modulated (PWM) dimming. An input signal from DC up to 20kHz can be applied to the DIM_IN pin (see Typical Application) to pulse the LED string ON and OFF. It is recommended to use PWM dimming signals above 120Hz to avoid any recognizable flicker by the human eye. PWM dimming is the preferred way to dim an LED in order to prevent color/wavelength shifting. Color wavelength shifting will occur with analog dimming. By employing PWM Dimming the output current level remains constant during each DIM_IN pulse. The boost converter switches only when DIM_IN is high. Between DIM_IN pulses the output capacitors will slowly discharge. The higher the DIM_IN frequency the less the output capacitors will discharge. PWM Dimming Limits The minimum pulse width of the DIM_IN is determined by the DIM_IN frequency and the L and C used in the boost stage output filter. At low DIM_IN frequencies lower dimming ratios can be achieved.
Dim_ratio = LED_ON_TIM E PERIOD PWMD
Figure 3. PWM Dimming 20%
Figure 3 shows that switching occurs only during DIM_IN on pulses. When DIM_IN is low the boost converter stops switching and the external LED is turned off. The LED current flows only when DIM_IN is high. Figure 3 shows that the compensation pin (VCOMP) does not discharge between DIM_IN pulses. Therefore, when the DIM_IN pulse starts again the converter resumes operation at the same VCOMP voltage. This eliminates the need for the comp pin to charge up during each DIM_IN pulse and allows for high Dimming ratios.
Figure 4. PWM Dimming 10% and ILED 100Hz Figure 2. DIM_IN Dimming Ratio
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Figure 5. PWM Dimming 20% and ILED 1kHz
Figure 7. 5s DIM_IN Pulse
In Figure 4 is at 100Hz dimming frequency and Figure 5 is 1kHz dimming frequency. The time it takes for the LED current to reach it full value is longer with a lower Dimming frequency. The reason is the output capacitors slowly discharge between dimming pulses.
Figure 7 shows the minimum DIM_IN pulse at these operating conditions before the ILED current starts to drop due to low VOUT. The converter is ON (switching) only during a DIM_IN pulse. Figure 7 shows that at this DIM_IN pulse width the converter is ON (switching) long enough to generate the necessary VOUT to forward bias the LED string at the programmed current level. Therefore this condition will result in the desired ILED.
Figure 6. PWM Dimming 20% and ILED 1kHz
Figure 6 shows the output voltage VOUT discharge between DIM_IN pulses. The amount of discharge is dependent on the time between DIM_IN pulses.
Figure 8. 2.5s DIM_IN Pulse
Figure 8 shows that at this DIM_IN pulse width the converter in not ON (switching) long enough to generate the necessary VOUT to forward bias the LED string at the programmed current level. As a result the LED current drops. Therefore, this condition will not result in the desired ILED.
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Design Procedure for a LED Driver
Symbol Input VIN IIN Output LEDs VF VOUT ILED IPP Pout DIM_IN OVP System FSW eff VDIODE Switching Frequency Efficiency Forward drop of schottky diode 1 80 0.5 MHz % V Number of LEDs Forward Voltage of LED Output Voltage LED Current Required I Ripple Output Power PWM Dimming Output Over Voltage Protection 0 30 5 3.2 16 0.33 6 3.5 21 0.35 40 10.36 100 7 4.0 28 0.37 V V A mA W % V Input Voltage Input Current 8 12 14 2 V A Parameter Min Nom Max Units
Table 1. Design example parameters
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Micrel, Inc. Design Example In this example, we will be designing a boost LED driver operating off a 12V input. This design has been created to drive 6 LEDs at 350mA with a ripple of about 20%. We are designing for 80% efficiency at a switching frequency of 1MHz. Select RADJ Having chosen the LED drive current to be 350mA in this example, the current can be set by choosing the RADJ resistor from Equation 1:
MIC3223
Using Equation 5, the following values have been calculated:
IIN_RMS(max) = VOUT(max) x IOUT(max) eff x VIN(min) VOUT(nom) x IOUT(nom) eff x VIN(nom) VOUT(min) x IOUT(min) eff x VIN(max) = 1.54A (RMS) = 0.74A (RMS) = 0.46A (RMS)
Eq (5)
IIN_RMS(nom ) = IIN_RMS(min) =
R ADJ =
0.2V = 0.57 0.35A
Use the next lowest standard value 0.56. ILED = 0.36A The power dissipation in this resistor is:
PRADJ = ILED 2 x R ADJ = 71mW
Use a resistor rated at quarter watt or higher.
Operating Duty Cycle The operating duty cycle can be calculated using Equation four provided below:
IOUT is the same as ILED. Selecting the inductor current (peak-to-peak), IL_PP, to be between 20% to 50% of IIN_RMS(nom), in this case 40%, we obtain: IIN_PP(nom) = 0.4 x IIN_RMS(nom) = 0.4 x 0.74 = 0.30AP-P It can be difficult to find large inductor values with high saturation currents in a surface mount package. Due to this, the percentage of the ripple current may be limited by the available inductor. It is recommended to operate in the continuous conduction mode. The selection of L described here is for continuous conduction mode. Eq. (6)
L= VIN x D I xF IN_PP SW
Eq. (4)
(V - VIN + VDIODE ) D = OUT
VOUT + VDIODE
VDIODE is the Vf of the output diode D1 in the Typical Application. It is recommended to use a schottky diode because it has a lower Vf than a junction diode. These can be calculated for the nominal (typical) operating conditions, but should also be understood for the minimum and maximum system conditions as listed below.
Dnom = Dmax = Dmin =
Using the nominal values, we get:
L= 12V x 0.44 0.3A x 1MHz = 18H
Select the next higher standard inductor value of 22H. Going back and calculating the actual ripple current gives:
IIN_PP(max) = VIN(min) x D max L x FSW = 8V x 0.72 = 0.26A PP 22 H x 1MHz
(VOUT(nom) - VIN(nom) + VDIODE )
(VOUT(max) - VIN(min) + VDIODE )
(VOUT(min) - VIN(max) + VDIODE )
VOUT(min) + VDIODE VOUT(max) + VDIODE
VOUT(nom) + VDIODE
The average input current is different than the RMS input current because of the ripple current. If the ripple current is low, then the average input current nearly equals the RMS input current. In the case where the average input current is different than the RMS, equation 7 shows the following: Eq. (7)
IIN_AVE(max) =
(21 - 12 - 0.5) = 0.44 Dnom =
21 + 0.5
21+ 0.5 Dnom
(IIN_RMS(max) )2 -
(IIN_PP ) 2 12
( 21- 12 + 0.5) = 0.44 =
IIN_AVE(max) = (1.54 )2 -
(0.24) 2 1.54 A 12
Therefore Dnom = 44%, Dmax = 72% and Dmin = 15%. Inductor Selection First calculate the RMS input current (nominal, min and max) for the system given the operating conditions listed in the design example table. The minimum value of the RMS input current is necessary to ensure proper operation.
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The Maximum Peak input current IL_PK can found using Equation 8: Eq. (8) IL_PK(max) = IIN_AVE(max) + 0.5 xIL_PP(max) = 1.67A The saturation current (ISAT) at the highest operating temperature of the inductor must be rated higher than this. The power dissipated in the inductor is:
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Micrel, Inc. Eq. (9) PINDUCTOR = IIN_RMS(max)2 x DCR A Coilcraft # MSS1260-223ML is used in this example. Its DCR is 52m, ISAT =2.7A PINDUCTOR = 1.542 x 52 m = 0.123W
Output Capacitor In this LED driver application, the ILED ripple current is a more important factor when compared to that of the output ripple voltage (although the two are directly related). To find the COUT for a required ILED ripple use the following calculation: For an output ripple ILED(ripple) = 20ma
IIN_PP VIN(ripple) x FSW
MIC3223
(0.3A) = 0.75 F 8 x 50mV x 1MHz
C IN =
=
This is the minimum value that should be used. To protect the IC from inductive spikes or any overshoot, a larger value of input capacitance may be required. Use 2.2F or higher as a good safe min.
Rectifier Diode Selection A schottky diode is best used here because of the lower forward voltage and the low reverse recovery time. The voltage stress on the diode is the max VOUT and therefore a diode with a higher rating than max VOUT should be used. An 80% de-rating is recommended here as well. Eq. (14) IDIODE(max) = IOUT(max) = 0.36A Since IIN_AVE(max) occurs when D is at a maximum. Eq. (15) PDIODE(max) VDIODE x IDIODE_(max) A SK35B is used in this example, it's VDIODE is 0.5V PDIODE(max) 0.5V x 0.36A = 0.18W MIC3223 Power Losses To find the power losses in the MIC3223: There is about 6mA input from VIN into the VDD pin. The internal power switch has an RDSON of about 170m at. PMIC3223 = VIN x 6mA + PwrFET Eq. (16) PwrFET = IFET_RMS(max)2 x Rds_on_@100 + VOUT(max) x IIN_AVE(max) x tsw x Fsw Rds_on_@100 160m tsw 30ns is the internal Power FET ON an OFF transition time.
2 IL_PP 2 ISWRMS(max) = D IIN_AVE(max) + 12
Eq. (10)
C OUT =
ILED(nom) x D nom ILED(ripple) x (R ADJ + R LED_total ) x FSW
Find the equivalent ac resistance RLED_ac from the datasheet of the LED. This is the inverse slope of the ILED vs. Vf curve i.e.: Eq. (11)
Vf LED In this example use RLED_ac = 0.6 for each LED. If the LEDs are connected in series, multiply RLED_ac = 0.6 by the total number of LEDs. In this example of six LEDs, we obtain the following: RLED_total Rdynamic = 6 x 0.6 = 3.6 Eq. (12) RLED_ac = C OUT = ILED(nom) x D nom ILED(ripple) x (R ADJ + R LED_total ) x FSW = 1.9 F
Use 2.2F or higher. There is a trade off between the output ripple and the rising edge of the DIM_IN pulse. This is because between PWM dimming pulses, the converter stops pulsing and COUT will start to discharge. The amount that COUT will discharge depends on the time between PWM Dimming pluses. At the next DIM_IN pulse, COUT has to be charged up to the full output voltage VOUT before the desired LED current flows.
Input Capacitor The input capacitor is shown in the Typical Application. For superior performance, ceramic capacitors should be used because of their low equivalent series resistance (ESR). The input capacitor CIN ripple current is equal to the ripple in the inductor. The ripple voltage across the input capacitor, CIN is the ESR of CIN times the inductor ripple. The input capacitor will also bypass the EMI generated by the converter as well as any voltage spikes generated by the inductance of the input line. For a required VIN(ripple):
= 1.3A
PwrFET = 1.3A2 x 160m + 28V x 1.54A x 30ns x 1MHz = 1.6W PMIC3223 = 8 x 6mA + 1.77W = 1.66W
Snubber A snubber is a damping resistor in series with a DC blocking capacitor in parallel with the power switch (same as across the flyback diode because VOUT is an ac ground). When the power switch turns off, the drain to source capacitance and parasitic inductance will cause a high frequency ringing at the switch node. A snubber circuit as shown in the application schematic may be required if ringing is present at the switch node. A critically damped circuit at the switch node is where R equals the characteristic impedance of the switch node.
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Lparisitic Cds
MIC3223
Eq.(17)
R
snubber
=
The explanation of the method to find the best R snubber is beyond the scope of this data sheet. Use Rsnubber = 2, 1/2 watt and Csnubber = 470pf to 1000pf. The power dissipation in the Rsnubber is: Rsnubber = Csnubber x VOUT2 x FSW Psnubber = 470pF x 28V2 x 1MHz = 0.4W
Power Loss in the L Power Loss in the sckottky diode Psnubber MIC3223 Power Loss Total Losses Efficiency Table 2. Major Power Losses 0.123 W 0.2 W 0.4 W 1.66 W 2.4W 80%
Figure 10. Simplified Control Loop
Eq. (19) Where
T(s) = Gea(s) x Gvc(s) x H(s)
For a LED driver H(s) =
R ADJ and R ADJ + R dynamic

1 Gea (s) = gm Z O || R comp + sCcomp Eq. (20)
G VC (s) = VOUT (s) VCONTROL (s)
Table 2 showing the Power losses in the Design Example.
OVP - Over Voltage Protection Set OVP higher than the maximum output voltage by at least one Volt. To find the resistor divider values for OVP use equation 18 and set the OVP = 30V and ROVP_H = 100k:
sL 1 - (1 + sC OUTRESR ) 2 1 D' R OP D' R dynamic = sR dynamic COUT Ri 2 1 + 2
Where
VOUT Is the DC operating point of the converter. ILED Rdymanic is the ac load the converter sees. When the load on the converter is a string of LEDs, Rdymanic is the series sum of the RLED(ac) of each LED. RLED_total is usually between 0.1 to 1 per LED. It can be calculated from the slope of ILED vs. Vf plot of the LED. Ri = Ai x Rcs = 0.86 Ai = 114 and Rcs 7.5m; are internal to the ic. The equation for Gvc(s) is theoretical and should give a good idea of where the poles and zeros are located. R OP =
Eq. (18)
Compensation
R OVP_L
100k x 1.245 = 4.33k = 30 - 1.245
Figure 9. Current Mode Loop Diagram
Current mode control simplifies the compensation. In current mode, the complex poles created by the output L and C are reduced to a single pole. The explanation for this is beyond the scope of this datasheet, but it's generally thought to be because the inductor becomes a constant current source and can't act to change phase. From the small signal block diagram the loop transfer function is:
Eq.(20) shows that s =
D'2 R dynamic
L 2L is a RHP Zero. The loop bandwidth should be about 1/5 to 1/10 of the frequency of RHPZ to ensure stability. From Equation (20) it is shown that there is only the single pole. 1 1 fpole = and a Zero s= R dynamic COUT 2R dynamic COUT
fRHPZ =
D'2 R dynamic
due to the ESR of the output capacitor.
s= 1 1 fESR = RESR COUT 2RESR COUT
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Micrel, Inc. This greatly simplifies the compensation. One needs only to get a bode plot of the transfer function of the control to output Gvc(s) with a network analyzer and/or calculate it. From the bode plot find what the gain of R Gvc(s) is at f = HPZ . Next design the error amp gain 10 Gea(s) so the loop gain at the cross over frequency T(fco) is R 0 db where fco = HPZ or less. 10
Error Amp
60 40 GAIN (dB) / PHASE () 20 0 -20 -40 -60 -80
MIC3223 The error amp is a gm type and the gain Gea(s) is Eq. (21)
gm =
1 Gea (s) = gm Z O || R comp + sCcomp

0.8mA and Zo = 1.2M. V f 1 R = co = HPZ . 2R compCcomp 10 100
The zero is fzero =
Error Amp Gain and Phase
Gain
Phase
1.E+02
1.E+03
1.E+04
1.E+05
1.E+06
FREQUENCY (Hz)
Figure 11. Internal Error Amp and External Compensation
Set the fco at the mid band where Gea(fco) = gm x Rcomp. At fzero x 10 the phase boost is near its maximum.
Figure 12. Error Amp Transfer Function
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Other Applications
Figure 13. MIC3223 Typical Application without External PWM Dimming Switch
Audio noise Audio noise from the output capacitors may exits in a standard boost LED converter. The physical dimensions of ceramic capacitors change with the voltage applied to them. During PWM Dimming, the output capacitors in standard converters are subjected to fast voltage and current transients that may cause the output capacitors to oscillate at the PWM Dimming frequency. This is one reason users may want PWM dimming frequencies above the audio range. PCB Layout 1. All typologies of DC-to-DC converters have a Reverse Recovery Current (RRC) of the flyback or (freewheeling) diode. Even a Schottky diode, which is advertised as having zero RRC, it really is not zero. The RRC of the freewheeling diode in a boost converter is even greater than in the Buck converter. This is because the output voltage is higher than the input voltage and the diode has to charge up to -VOUT during each ontime pulse and then discharge to Vf during the off-time.
2. Even though the RRC is very short (tens of nanoseconds) the peak currents are high (multiple amperes). These fast current spikes generate EMI (electromagnetic interference). The amount of RRC is related to the die size and internal capacitance of the diode. It is important not to oversize (i.e. not more than the usual de rating) the diode because the RRC will be needlessly higher. Example: If a 2A diode is needed do not use a higher current rated diode because the RRC will be needlessly higher. If a 25V diode is needed do not use a 100V etc. 3. The high RRC causes a voltage drop on the ground trace of the PCB and if the converter control IC is referenced to this voltage drop, the output regulation will suffer. 4. For good output regulation, it is important to connect the IC's reference to the same point as the output capacitors to avoid the voltage drop caused by RRC. This is also called a star connection or single point grounding. 5. Feedback trace: The high impedance traces of the FB should be short.
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Evaluation Board Schematic
37V Max 1A LED Driver
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Bill of Materials
Item C1 C2 C3, C7 Part Number GRM319R61E475KA12D C3216X7R1E475M 12063D475KAT2A GRM188R71C273KA01D GRM188R60J106ME47D C1608X5R0J106K 08056D106MAT2A C4, C6 C5 C8 D1 L1 R1, R3 R2 R4 R5 R6 Q1 U1
Notes: 1. Murata: www.murata.com. 2. TDK: www.tdk.com. 3. AVX: www.avx.com. 4. Vishay: www.vishay.com. 5. Internacional Rectifier: www.ift.com. 6. Coilcraft: www.coilcraft.com 7. Stackpole Electronics, Inc.: www. 8. Analog Power: www.analogpowerinc.com 8. Micrel, Inc.: www.micrel.com.
Manufacturer muRata
(1)
Description Ceramic Capacitor, 4.7F, 25V, Size 1206, X7R Ceramic Capacitor, 0.027F, 6.3V, Size 0603, X7R Ceramic Capacitor, 10F, 6.3V, Size 0603, X7R
Qty 1 1 2
TDK(2) AVX(3) muRata muRata TDK AVX AVX muRata muRata AVX muRata MCC
(4)
12105C475KAZ2A GRM32ER71H475KA88L GRM188R71C473KA01D 0603YC473K4T2A GRM188R72A102KA37D SK35B MSD1260-223ML-LD CRCW0603100KFKEA CRCW0603549RFKEA CRCW06033K24FKEA CRCW1206R560FKEA RMC 1/4 2 1% R Si2318DS AM2340N MIC3223
Ceramic Capacitor, 4.7F, 50V, Size 1210, X7R Ceramic Capacitor, 0.047F, 6.3V, Size 0603, X7R Ceramic Capacitor, 1000pF, 100V Size 0603, X7R Schottky Diode, 3A, 50V (SMB) Inductor, 22H, 5A Resistor, 100k, 1%, Size 0603 Resistor, 549, 1%, Size 0603 Resistor, 3.24k, 1%, Size 0603 Resistor, 0.56, 1%, 1/2W, Size 1206 (for .35A LED current Change for different ILED) Resistor, 2, 1%, 1/2W, Size 1210 N-Channel 40V MOSFET High Power Boost LED Driver with Integrated FET
2 1
1 1 2 1 1 1 1 1 1
Coilcraft(6) Vishay Dale(4) Vishay Dale Vishay Dale Vishay Dale Stackpole Electronics, Inc.(7) Vishay Siliconix(4) Analog Power
(8)
Micrel, Inc.(9)
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MIC3223
PCB Layout Recommendations
Top Layer
Bottom Layer
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Package Information
16-Pin ePad TSSOP (TSE)
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Recommended Land Pattern
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2009 Micrel, Incorporated.
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