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ISL6323A
Data Sheet March 23, 2009 FN6878.0
Monolithic Dual PWM Hybrid Controller Powering AMD SVI Split-Plane and PVI Uniplane Processors
The ISL6323A dual PWM controller delivers high efficiency and tight regulation from two synchronous buck DC/DC converters. The ISL6323A supports hybrid power control of AMD processors which operate from either a 6-bit parallel VID interface (PVI) or a serial VID interface (SVI). The dual output ISL6323A features a multi-phase controller to support uniplane VDD core voltage and a single phase controller to power the Northbridge (VDDNB) in SVI mode. Only the multi-phase controller is active in PVI mode to support uniplane VDD only processors. A precision uniplane core voltage regulation system is provided by a two-to-four-phase PWM voltage regulator (VR) controller. The integration of two power MOSFET drivers, adding flexibility in layout, reduce the number of external components in the multi-phase section. A single phase PWM controller with integrated driver provides a second precision voltage regulation system for the North Bridge portion of the processor. This monolithic, dual controller with integrated driver solution provides a cost and space saving power management solution. For applications which benefit from load line programming to reduce bulk output capacitors, the ISL6323A features output voltage droop. The multi-phase portion also includes advanced control loop features for optimal transient response to load apply and removal. One of these features is highly accurate, fully differential, continuous DCR current sensing for load line programming and channel current balance. Dual edge modulation is another unique feature, allowing for quicker initial response to high di/dt load transients. The ISL6323A supports Power Savings Mode by dropping the number of phases to one when the PSI_L bit is set.
Features
* Processor Core Voltage Via Integrated Multi-Phase Power Conversion * Configuration Flexibility - 2-Phase Operation with Internal Drivers - 3- or 4-Phase Operation with External PWM Drivers * PSI_L Support with Phase Shedding for Improved Efficiency at Light Load * Serial VID Interface Inputs - Two Wire, Clock and Data, Bus - Conforms to AMD SVI Specifications * Parallel VID Interface Inputs - 6-bit VID input - 0.775V to 1.55V in 25mV Steps - 0.375V to 0.7625V in 12.5mV Steps * Precision Core Voltage Regulation - Differential Remote Voltage Sensing - 0.6% System Accuracy Over-Temperature - Adjustable Reference-Voltage Offset * Optimal Processor Core Voltage Transient Response - Adaptive Phase Alignment (APA) - Active Pulse Positioning Modulation * Fully Differential, Continuous DCR Current Sensing - Accurate Load Line Programming - Precision Channel Current Balancing * Variable Gate Drive Bias: 5V to 12V * Overcurrent Protection * Multi-tiered Overvoltage Protection * Selectable Switching Frequency up to 1MHz * Simultaneous Digital Soft-Start of Both Outputs * Processor NorthBridge Voltage Via Single Phase Power Conversion * Precision Voltage Regulation - Differential Remote Voltage Sensing - 0.6% System Accuracy Over Temperature * Serial VID Interface Inputs - Two Wire, Clock and Data, Bus - Conforms to AMD SVI Specifications * Overcurrent Protection * Continuous DCR Current Sensing * Variable Gate Drive Bias: 5V to 12V * Simultaneous Digital Soft-Start of Both Outputs * Selectable Switching Frequency up to 1MHz * Pb-Free Plus Anneal Available (RoHS Compliant)
Ordering Information
PART NUMBER (Note) ISL6323ACRZ* ISL6323AIRZ* PART MARKING TEMP. (C) PACKAGE (Pb-free) PKG. DWG. #
ISL6323A CRZ 0 to +70 48 Ld 7x7 QFN L48.7x7 ISL6323A IRZ -40 to +85 48 Ld 7x7 QFN L48.7x7
*Add "-T" suffix for tape and reel. Please refer to TB347 for details on reel specifications. NOTE: These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2009. All Rights Reserved All other trademarks mentioned are the property of their respective owners.
ISL6323A Pinout
ISL6323A (48 LD QFN) TOP VIEW
VDDPWRGD 37 36 PWM4 35 PWM3 34 PWROK 33 PHASE1 32 UGATE1 49 GND 31 BOOT1 30 LGATE1 29 PVCC1_2 28 LGATE2 27 BOOT2 26 UGATE2 25 PHASE2 13 VSEN 14 OFS 15 DVC 16 RSET 17 FB 18 COMP 19 APA 20 ISEN1+ 21 ISEN122 ISEN2+ 23 ISEN224 EN PHASE_NB 38 UGATE_NB 39 LGATE_NB 41 COMP_NB BOOT_NB 40 PVCC_NB 42 ISEN_NB-
ISEN4+
ISEN3+ 44
ISEN4-
48 FB_NB ISEN_NB+ RGND_NB VID0/VFIXEN VID1/SEL VID2/SVD VID3/SVC VID4 VID5 1 2 3 4 5 6 7 8 9
47
46
45
43
VCC 10 FS 11 RGND 12
Integrated Driver Block Diagram
ISEN3-
PVCC BOOT
UGATE PWM 20K SOFT-START AND FAULT LOGIC SHOOTTHROUGH PROTECTION 10K
GATE CONTROL LOGIC
PHASE
LGATE
2
FN6878.0
ISL6323A Controller Block Diagram
RGND_NB FB_NB COMP_NB
NB_REF
BOOT_NB E/A MOSFET DRIVER RAMP UGATE_NB PHASE_NB LGATE_NB
ISEN_NB+ ISEN_NB-
CURRENT SENSE
UV LOGIC
OV LOGIC
VDDPWRGD APA COMP APA NB FAULT LOGIC
EN_12V PVCC_NB ENABLE LOGIC POWER-ON RESET SOFT-START AND E/A FAULT LOGIC BOOT1 DROOP CONTROL LOAD APPLY TRANSIENT ENHANCEMENT SVI SLAVE BUS AND PVI DAC UGATE1 PHASE1 LGATE1 CLOCK AND TRIANGLE WAVE GENERATOR 2X EN VCC PVCC1_2
OFS
OFFSET
FB DVC RGND
MOSFET DRIVER
PWROK VID0/VFIXEN VID1/SEL VID2/SVD VID3/SVC VID4 VID5
FS
PWM1
NB_REF OV LOGIC VSEN UV LOGIC
RESISTOR MATCHING
BOOT2
PWM2
PWM3
MOSFET DRIVER
UGATE2 PHASE2 LGATE2
OC
PWM4 PH3/PH4 POR
RSET
ISEN1+ ISEN1ISEN2+ ISEN2ISEN3+ ISEN3ISEN4+ ISEN4-
CH1 CURRENT SENSE
I_TRIP
I_AVG
CHANNEL DETECT
EN_12V
ISEN3ISEN4-
CH2 CURRENT SENSE
CHANNEL CURRENT BALANCE
I_AVG
1 N
PWM3 SIGNAL LOGIC
CH3 CURRENT SENSE ISEN3CH4 CURRENT SENSE ISEN4-
PWM3
PWM4 SIGNAL LOGIC
PWM4
GND
3
FN6878.0
ISL6323A Typical Application - SVI Mode
+12V FB COMP ISEN3+ ISEN3PWM3 VSEN BOOT1 UGATE1 PHASE1
+12V
BOOT1 UGATE1 PHASE1
LGATE1 LGATE1 APA DVC ISEN1ISEN1+ PGND
PWM1
ISL6614 +12V +5V VCC BOOT2 OFS FS +5V UGATE2 PHASE2 CPU LOAD UGATE2GND PHASE2 PWM2 LGATE2 PVCC1_2 VDD +12V
+12V
VCC BOOT2 PVCC
RSET
LGATE2
NC NC
VFIXEN ISEN2SEL ISEN2+ SVD SVC RGND VID4 VID5 PWROK VDDPWRGD ISEN4+ GND ISEN4PWM4
+12V
ISL6323A
+12V PVCC_NB EN BOOT_NB UGATE_NB PHASE_NB VDDNB
OFF ON
LGATE_NB COMP_NB ISEN_NBISEN_NB+ FB_NB NB LOAD
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ISL6323A Typical Application - PVI Mode
+12V FB COMP ISEN3+ ISEN3PWM3 VSEN BOOT1 UGATE1 PHASE1
+12V
BOOT1 UGATE1 PHASE1
LGATE1 APA LGATE1 ISEN1ISEN1+ +12V +5V VCC BOOT2 OFS +5V FS UGATE2 PHASE2 CPU LOAD LGATE2 PVCC1_2 VDD PGND
PWM1
DVC
ISL6614
+12V VCC BOOT2 PVCC
+12V
GND UGATE2 PHASE2 PWM2
RSET
LGATE2
NC
VID0 ISEN2VID1/SEL ISEN2+ VID2 VID3 RGND VID4 VID5 PWROK VDDPWRGD ISEN4+ GND ISEN4PWM4
+12V
ISL6323A +12V PVCC_NB NORTH BRIDGE REGULATOR DISABLED IN PVI MODE BOOT_NB UGATE_NB PHASE_NB VDDNB EN
OFF ON
LGATE_NB COMP_NB ISEN_NBISEN_NB+ FB_NB NB LOAD
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ISL6323A
Absolute Maximum Ratings
Supply Voltage (VCC) . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6.2V Supply Voltage (PVCC) . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +15V Absolute Boot Voltage (VBOOT). . . . . . . . GND - 0.3V to GND + 36V Phase Voltage (VPHASE) . . . . . . . . GND - 0.3V to 15V (PVCC = 12) GND - 8V (<400ns, 20J) to 24V (<200ns, VBOOT-PHASE = 12V) Upper Gate Voltage (VUGATE). . . . VPHASE - 0.3V to VBOOT + 0.3V VPHASE - 3.5V (<100ns Pulse Width, 2J) to VBOOT + 0.3V Lower Gate Voltage (VLGATE) . . . . . . . GND - 0.3V to PVCC + 0.3V GND - 5V (<100ns Pulse Width, 2J) to PVCC+ 0.3V Input, Output, or I/O Voltage . . . . . . . . . GND - 0.3V to VCC + 0.3V
Thermal Information
Thermal Resistance JA (C/W) JC (C/W) QFN Package (Notes 1, 2) . . . . . . . . . . 27 2 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +150C Maximum Storage Temperature Range . . . . . . . . . .-65C to +150C Pb-free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . .see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
VCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+5V 5% PVCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . +5V to 12V 5% Ambient Temperature ISL6323ACRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0C to +70C ISL6323AIRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40C to +85C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty.
NOTES: 1. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with "direct attach" features. See Tech Brief TB379. 2. For JC, the "case temp" location is the center of the exposed metal pad on the package underside. 3. Limits established by characterization and are not production tested.
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Specified. Parameters with MIN and/or MAX limits are 100% tested at +25C, unless otherwise specified. Temperature limits established by characterization and are not production tested. TEST CONDITIONS MIN TYP MAX UNITS
PARAMETER BIAS SUPPLIES Input Bias Supply Current Gate Drive Bias Current - PVCC1_2 Pin Gate Drive Bias Current - PVCC_NB Pin VCC POR (Power-On Reset) Threshold
IVCC; EN = high IPVCC1_2; EN = high IPVCC_NB; EN = high VCC Rising VCC Falling
15 1 0.3 4.20 3.70 4.20 3.70
22 1.8 0.9 4.35 3.85 4.35 3.85
25 3 2 4.50 4.05 4.50 4.05
mA mA mA V V V V
PVCC POR (Power-On Reset) Threshold
PVCC Rising PVCC Falling
PWM MODULATOR Oscillator Frequency Accuracy, FSW RT = 100k (0.1%) to Ground, (Droop Enabled) RT = 100k (0.1%) to VCC, (Droop Disabled), 0C to +70C RT = 100k (0.1%) to VCC, (Droop Disabled), -40C to +85C Typical Adjustment Range of Switching Frequency Oscillator Ramp Amplitude, VP-P Maximum Duty Cycle CONTROL THRESHOLDS EN Rising Threshold EN Hysteresis PWROK Input HIGH Threshold PWROK Input LOW Threshold 0.80 70 0.88 130 1.1 0.95 0.92 190 V mV V V (Note 3) (Note 3) (Note 3) 225 245 240 0.08 1.50 99.5 250 275 275 265 310 310 1.0 kHz kHz kHz MHz V %
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Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Specified. Parameters with MIN and/or MAX limits are 100% tested at +25C, unless otherwise specified. Temperature limits established by characterization and are not production tested. (Continued) TEST CONDITIONS Open drain, V_VDDPWRGD = 400mV VISEN3-, VISEN4-, VISEN24.4 MIN TYP MAX 4 UNITS mA V
PARAMETER VDDPWRGD Sink Current PWM Channel Disable Threshold PIN_ADJUSTABLE OFFSET OFS Source Current Accuracy (Positive Offset) OFS Sink Current Accuracy (Negative Offset) REFERENCE AND DAC System Accuracy (VDAC > 1.000V) System Accuracy (0.600V < VDAC < 1.000V) System Accuracy (VDAC < 0.600V) DVC Voltage Gain APA Current Tolerance ERROR AMPLIFIER DC Gain Gain-Bandwidth Product (Note 3) Slew Rate (Note 3) Maximum Output Voltage Minimum Output Voltage SOFT-START RAMP Soft-Start Ramp Rate PWM OUTPUTS PWM Output Voltage LOW Threshold PWM Output Voltage HIGH Threshold CURRENT SENSING - CORE CONTROLLER Sensed Current Tolerance CURRENT SENSING - NB CONTROLLER Sensed Current Tolerance DROOP CURRENT Tolerance OVERCURRENT PROTECTION Overcurrent Trip Level - Average Channel VDAC = 1V VAPA = 1V
ROFS = 10k ( 0.1%) from OFS to GND ROFS = 30k ( 0.1%) from OFS to VCC
27.5 50.5
31 53.5
34.5 56.5
A A
-0.6 -1.0 -2.0 2.0 90 100
0.6 1.0 2.0
% % % V
108
A
RL = 10k to ground, (Note 3) CL = 100pF, RL = 10k to ground, (Note 3) CL = 100pF, Load = 400A, (Note 3) Load = 1mA Load = -1mA 3.80
96 20 8 4.20 1.3 1.6
dB MHz V/s V V
2.2
3.0
4.0
mV/s
ILOAD = 500A ILOAD = 500A 4.5
0.5
V V
VISENn- - VISENn+ = 23.2mV, RSET = 37.6k, 4 Phases, TA = +25C
68
88
A
VISEN_NB- - VISEN_NB+ = 23.2mV, RSET = 37.6k, 4 Phases, TA = +25C
68
89
A
VISENn- - VISENn+ = 23.2mV, RSET = 37.6k, 4 Phases, TA = +25C
68
88
A
Normal Operation, RSET = 28.2k Dynamic VID Change (Note 3)
87
100 130 142 190
120
A A A A
Overcurrent Limiting - Individual Channel
Normal Operation, RSET = 28.2k Dynamic VID Change (Note 3)
POWER GOOD Core Overvoltage Threshold VSEN Rising VDAC +225mV VDAC + 250mV VDAC +275mV
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ISL6323A
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Specified. Parameters with MIN and/or MAX limits are 100% tested at +25C, unless otherwise specified. Temperature limits established by characterization and are not production tested. (Continued) TEST CONDITIONS VSEN Falling ISEN_NB+ Falling MIN VDAC -325mV VDAC -310mV TYP VDAC -300mV VDAC -275mV 50 MAX VDAC -270mV VDAC -235mV mV UNITS
PARAMETER Core Undervoltage Threshold NB Undervoltage Threshold Power Good Hysteresis OVERVOLTAGE PROTECTION OVP Trip Level OVP Lower Gate Release Threshold SWITCHING TIME (Note 3) [See "Timing Diagram" on page 9] UGATE Rise Time LGATE Rise Time UGATE Fall Time LGATE Fall Time UGATE Turn-On Non-overlap LGATE Turn-On Non-overlap GATE DRIVE RESISTANCE (Note 3) Upper Drive Source Resistance Upper Drive Sink Resistance Lower Drive Source Resistance Lower Drive Sink Resistance MODE SELECTION VID1/SEL Input Low VID1/SEL Input High PVI INTERFACE VIDx Pull-down VIDx Input Low VIDx Input High SVI INTERFACE SVC, SVD Input HIGH (VIH) SVC, SVD Input LOW (VIL) Schmitt Trigger Input Hysteresis SVD Low Level Output Voltage Maximum SVC, SVD Leakage (Note 3)
1.73 350
1.80 400
1.84
V mV
tRUGATE; VPVCC = 12V, 3nF Load, 10% to 90% tRLGATE; VPVCC = 12V, 3nF Load, 10% to 90% tFUGATE; VPVCC = 12V, 3nF Load, 90% to 10% tFLGATE; VPVCC = 12V, 3nF Load, 90% to 10% tPDHUGATE; VPVCC = 12V, 3nF Load, Adaptive tPDHLGATE; VPVCC = 12V, 3nF Load, Adaptive
26 18 18 12 10 10
ns ns ns ns ns ns
VPVCC = 12V, 15mA Source Current VPVCC = 12V, 15mA Sink Current VPVCC = 12V, 15mA Source Current VPVCC = 12V, 15mA Sink Current
2.0 1.65 1.25 0.80

EN taken from LO to HI, VDDIO = 1.5V EN taken from LO to HI, VDDIO = 1.5V 1.00
0.6
V V
VDDIO = 1.5V VDDIO = 1.5V VDDIO = 1.5V 1.00
30
40 0.6
A V V
0.95 0.4 0.14 3mA Sink Current 5 0.35 0.45 0.285
V V V V A
8
FN6878.0
ISL6323A Timing Diagram
tPDHUGATE tRUGATE UGATE LGATE tFUGATE
tFLGATE tPDHLGATE
tRLGATE
Functional Pin Description
VID1/SEL
This pin selects SVI or PVI mode operation based on the state of the pin prior to enabling the ISL6323A. If the pin is LO prior to enable, the ISL6323A is in SVI mode and the dual purpose pins [VID0/VFIXEN, VID2/SVC, VID3/SVD] use their SVI mode related functions. If the pin held HI prior to enable, the ISL6323A is in PVI mode and dual purpose pins use their VIDx related functions to decode the correct DAC code.
VID5
This pin is active only when the ISL6323A is in PVI mode. When VID1 is HI prior to enable, the ISL6323A decodes the programmed DAC voltage required by the AMD processor. This pin has an internal 30A pull-down current applied to it at all times.
VCC
VCC is the bias supply for the ICs small-signal circuitry. Connect this pin to a +5V supply and decouple using a quality 0.1F ceramic capacitor.
VID0/VFIXEN
If VID1 is LO prior to enable [SVI Mode], the pin is functions as the VFIXEN selection input from the AMD processor for determining SVI mode versus VFIX mode of operation. If VID1 is HI prior to enable [PVI Mode], the pin is used as DAC input VID0. This pin has an internal 30A pull-down current applied to it at all times.
PVCC1_2
The power supply pin for the multi-phase internal MOSFET drivers. Connect this pin to any voltage from +5V to +12V depending on the desired MOSFET gate-drive level. Decouple this pin with a quality 1.0F ceramic capacitor.
PVCC_NB
The power supply pin for the internal MOSFET driver for the Northbridge controller. Connect this pin to any voltage from +5V to +12V depending on the desired MOSFET gate-drive level. Decouple this pin with a quality 1.0F ceramic capacitor.
VID2/SVD
If VID1 is LO prior to enable [SVI Mode], this pin is the serial VID data bi-directional signal to and from the master device on AMD processor. If VID1 is HI prior to enable [PVI Mode], this pin is used to decode the programmed DAC code for the processor. In PVI mode, this pin has an internal 30A pull-down current applied to it. There is no pulldown current in SVI mode.
GND
GND is the bias and reference ground for the IC. The GND connection for the ISL6323A is through the thermal pad on the bottom of the package.
VID3/SVC
If VID1 is LO prior to enable [SVI Mode], this pin is the serial VID clock input from the AMD processor. If VID1 is HI prior to enable [PVI Mode], the ISL6323A is in PVI mode and this pin is used to decode the programmed DAC code for the processor. In PVI mode, this pin has an internal 30A pull-down current applied to it. There is no pulldown current in SVI mode.
EN
This pin is a threshold-sensitive (approximately 0.85V) system enable input for the controller. Held low, this pin disables both CORE and NB controller operation. Pulled high, the pin enables both controllers for operation. When the EN pin is pulled high, the ISL6323A will be placed in either SVI or PVI mode. The mode is determined by the latched value of VID1 on the rising edge of the EN signal. A third function of this pin is to provide driver bias monitor for external drivers. A resistor divider with the center tap connected to this pin from the drive bias supply prevents enabling the controller before insufficient bias is provided to external driver. The resistors should be selected such that
VID4
This pin is active only when the ISL6323A is in PVI mode. When VID1 is HI prior to enable, the ISL6323A decodes the programmed DAC voltage required by the AMD processor. This pin has an internal 30A pull-down current applied to it at all times.
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FN6878.0
ISL6323A
when the POR-trip point of the external driver is reached, the voltage at this pin meets the above mentioned threshold level.
BOOT1 and BOOT2
These pins provide the bias voltage for the corresponding upper MOSFET drives. Connect these pins to appropriately-chosen external bootstrap capacitors. Internal bootstrap diodes connected to the PVCC1_2 pin provide the necessary bootstrap charge.
FS
A resistor, placed from FS to Ground or from FS to VCC, sets the switching frequency of both controllers. Refer to Equation 1 for proper resistor calculation.
R T = 10
[ 10.61 - 1.035 log ( f ) ] s
(EQ. 1)
PHASE1 and PHASE2
Connect these pins to the sources of the corresponding upper MOSFETs. These pins are the return path for the upper MOSFET drives.
With the resistor tied from FS to Ground, Droop is enabled. With the resistor tied from FS to VCC, Droop is disabled.
VSEN and RGND
VSEN and RGND are inputs to the core voltage regulator (VR) controller precision differential remote-sense amplifier and should be connected to the sense pins of the remote processor core(s), VDDFB[H,L].
LGATE1 and LGATE2
These pins are used to control the lower MOSFETs. Connect these pins to the corresponding lower MOSFETs' gates.
PWM3 and PWM4
Pulse-width modulation outputs. Connect these pins to the PWM input pins of an Intersil driver IC if 3- or 4-phase operation is desired. Connect the ISEN- pins of the channels not desired to +5V to disable them and configure the core VR controller for 2- or 3-phase operation.
FB and COMP
These pins are the internal error amplifier inverting input and output respectively of the core VR controller. FB, VSEN and COMP are tied together through external R-C networks to compensate the regulator.
PWROK
System wide Power Good signal. If this pin is low, the two SVI bits are decoded to determine the "metal VID". When pin is high, the SVI is actively running its protocol.
APA
Adaptive Phase Alignment (APA) pin for setting trip level and adjusting time constant. A 100A current flows into the APA pin and by tying a resistor from this pin to COMP the trip level for the Adaptive Phase Alignment circuitry can be set.
RSET
Connect this pin to VCC through a resistor to set the effective value of the internal RISEN current sense resistors. An external PTC thermistor network can also be used to thermally compensate the current sense resistors to account for changes in inductor DCR over-temperature.
OFS
The OFS pin provides a means to program a dc current for generating an offset voltage across the resistor between FB and VSEN The offset current is generated via an external resistor and precision internal voltage references. The polarity of the offset is selected by connecting the resistor to GND or VCC. For no offset, the OFS pin should be left unconnected.
VDDPWRGD
During normal operation this pin indicates whether both output voltages are within specified overvoltage and undervoltage limits. If either output voltage exceeds these limits or a reset event occurs (such as an overcurrent event), the pin is pulled low. This pin is always low prior to the end of soft-start.
ISEN1-, ISEN1+, ISEN2-, ISEN2+, ISEN3-, ISEN3+, ISEN4- and ISEN4+
These pins are used for differentially sensing the corresponding channel output currents. The sensed currents are used for channel balancing, protection, and core load line regulation. Connect ISEN1-, ISEN2-, ISEN3-, and ISEN4- to the node between the RC sense elements surrounding the inductor of their respective channel. Tie the ISEN+ pins to the VCORE side of their corresponding channel's sense capacitor.
RGND_NB
This pin is an input to the NB VR controller precision differential remote-sense amplifier and should be connected to the sense pin of the North Bridge, VDDNBFBL.
DVC
The DVC pin is a buffered version of the reference to the error amplifier. A series resistor and capacitor between the DVC pin and FB pin smooth the voltage transition during VID-on-the-fly operations.
UGATE1 and UGATE2
Connect these pins to the corresponding upper MOSFET gates. These pins are used to control the upper MOSFETs and are monitored for shoot-through prevention purposes. Maximum individual channel duty cycle is limited to 93.3%.
FB_NB and COMP_NB
These pins are the internal error amplifier inverting input and output respectively of the NB VR controller. FB_NB, VDIFF_NB, and COMP_NB are tied together through external R-C networks to compensate the regulator.
FN6878.0
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ISL6323A
ISEN_NB-, ISEN_NB+
These pins are used for differentially sensing the North Bridge output current. The sensed current is used for protection and load line regulation if droop is enabled. Connect ISEN_NB- to the node between the RC sense element surrounding the inductor. Tie the ISEN_NB+ pin to the VNB side of the sense capacitor. channels. In a 3-phase converter, each channel switches 1/3 cycle after the previous channel and 1/3 cycle before the following channel. As a result, the three-phase converter has a combined ripple frequency three times greater than the ripple frequency of any one phase. In addition, the peak-topeak amplitude of the combined inductor currents is reduced in proportion to the number of phases (Equations 2 and 3). Increased ripple frequency and lower ripple amplitude mean that the designer can use less per-channel inductance and lower total output capacitance for any performance specification. Figure 1 illustrates the multiplicative effect on output ripple frequency. The three channel currents (IL1, IL2, and IL3) combine to form the AC ripple current and the DC load current. The ripple component has three times the ripple frequency of each individual channel current. Each PWM pulse is terminated 1/3 of a cycle after the PWM pulse of the previous phase. The peak-to-peak current for each phase is about 7A, and the DC components of the inductor currents combine to feed the load.
UGATE_NB
Connect this pin to the corresponding upper MOSFET gate. This pin provides the PWM-controlled gate drive for the upper MOSFET and is monitored for shoot-through prevention purposes.
BOOT_NB
This pin provides the bias voltage for the corresponding upper MOSFET drive. Connect this pin to appropriately-chosen external bootstrap capacitor. The internal bootstrap diode connected to the PVCC_NB pin provides the necessary bootstrap charge.
PHASE_NB
Connect this pin to the source of the corresponding upper MOSFET. This pin is the return path for the upper MOSFET drive. This pin is used to monitor the voltage drop across the upper MOSFET for overcurrent protection.
IL1 + IL2 + IL3, 7A/DIV
IL3, 7A/DIV PWM3, 5V/DIV IL2, 7A/DIV PWM2, 5V/DIV IL1, 7A/DIV PWM1, 5V/DIV 1s/DIV
LGATE_NB
Connect this pin to the corresponding MOSFET's gate. This pin provides the PWM-controlled gate drive for the lower MOSFET. This pin is also monitored by the adaptive shootthrough protection circuitry to determine when the lower MOSFET has turned off.
Operation
The ISL6323A utilizes a multi-phase architecture to provide a low cost, space saving power conversion solution for the processor core voltage. The controller also implements a simple single phase architecture to provide the Northbridge voltage on the same chip.
FIGURE 1. PWM AND INDUCTOR-CURRENT WAVEFORMS FOR 3-PHASE CONVERTER
Multi-phase Power Conversion
Microprocessor load current profiles have changed to the point that the advantages of multi-phase power conversion are impossible to ignore. The technical challenges associated with producing a single-phase converter that is both cost-effective and thermally viable have forced a change to the cost-saving approach of multi-phase. The ISL6323A controller helps simplify implementation by integrating vital functions and requiring minimal external components. The "Controller Block Diagram" on page 3 provides a top level view of the multi-phase power conversion using the ISL6323A controller.
To understand the reduction of ripple current amplitude in the multi-phase circuit, examine Equation 2, which represents an individual channel peak-to-peak inductor current.
( V IN - V OUT ) V OUT I PP = ----------------------------------------------------L fS V
IN
(EQ. 2)
In Equation 2, VIN and VOUT are the input and output voltages respectively, L is the single-channel inductor value, and fS is the switching frequency. The output capacitors conduct the ripple component of the inductor current. In the case of multi-phase converters, the capacitor current is the sum of the ripple currents from each of the individual channels. Compare Equation 2 to the expression for the peak-to-peak current after the summation of N symmetrically phase-shifted inductor currents in Equation 3. Peak-to-peak ripple current decreases by an amount proportional to the number of channels. Outputvoltage ripple is a function of capacitance, capacitor equivalent series resistance (ESR), and inductor ripple
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Interleaving
The switching of each channel in a multi-phase converter is timed to be symmetrically out of phase with each of the other 11
ISL6323A
current. Reducing the inductor ripple current allows the designer to use fewer or less costly output capacitors.
( V IN - N V OUT ) V OUT I C, PP = ----------------------------------------------------------L fS V
IN
(EQ. 3)
and turn off immediately after the PWM signal has transitioned high. This is important because it allows the controller to quickly respond to output voltage drops associated with current load spikes, while avoiding the ring back affects associated with other modulation schemes. The PWM output state is driven by the position of the error amplifier output signal, VCOMP, minus the current correction signal relative to the proprietary modulator ramp waveform as illustrated in Figure 3. At the beginning of each PWM time interval, this modified VCOMP signal is compared to the internal modulator waveform. As long as the modified VCOMP voltage is lower then the modulator waveform voltage, the PWM signal is commanded low. The internal MOSFET driver detects the low state of the PWM signal and turns off the upper MOSFET and turns on the lower synchronous MOSFET. When the modified VCOMP voltage crosses the modulator ramp, the PWM output transitions high, turning off the synchronous MOSFET and turning on the upper MOSFET. The PWM signal will remain high until the modified VCOMP voltage crosses the modulator ramp again. When this occurs the PWM signal will transition low again. During each PWM time interval the PWM signal can only transition high once. Once PWM transitions high it can not transition high again until the beginning of the next PWM time interval. This prevents the occurrence of double PWM pulses occurring during a single period. To further improve the transient response, ISL6323A also implements Intersil's proprietary Adaptive Phase Alignment (APA) technique, which turns on all phases together under transient events with large step current. With both APP and APA control, ISL6323A can achieve excellent transient performance and reduce the demand on the output capacitors.
Another benefit of interleaving is to reduce input ripple current. Input capacitance is determined in part by the maximum input ripple current. Multi-phase topologies can improve overall system cost and size by lowering input ripple current and allowing the designer to reduce the cost of input capacitance. The example in Figure 2 illustrates input currents from a three-phase converter combining to reduce the total input ripple current. The converter depicted in Figure 2 delivers 1.5V to a 36A load from a 12V input. The RMS input capacitor current is 5.9A. Compare this to a single-phase converter also stepping down 12V to 1.5V at 36A. The single-phase converter has 11.9ARMS input capacitor current. The single-phase converter must use an input capacitor bank with twice the RMS current capacity as the equivalent three-phase converter.
INPUT-CAPACITOR CURRENT, 10A/DIV
CHANNEL 3 INPUT CURRENT 10A/DIV
CHANNEL 2 INPUT CURRENT 10A/DIV
CHANNEL 1 INPUT CURRENT 10A/DIV 1s/DIV
FIGURE 2. CHANNEL INPUT CURRENTS AND INPUTCAPACITOR RMS CURRENT FOR 3-PHASE CONVERTER
Adaptive Phase Alignment (APA)
To further improve the transient response, the ISL6323A also implements Intersil's proprietary Adaptive Phase Alignment (APA) technique, which turns on all of the channels together at the same time during large current step transient events. As Figure 3 shows, the APA circuitry works by monitoring the voltage on the APA pin and comparing it to a filtered copy of the voltage on the COMP pin. The voltage on the APA pin is a copy of the COMP pin voltage that has been negatively offset. If the APA pin exceeds the filtered COMP pin voltage an APA event occurs and all of the channels are forced on.
Figures 25, 26 and 27 in the section entitled "Input Capacitor Selection" on page 32 can be used to determine the input-capacitor RMS current based on load current, duty cycle, and the number of channels. They are provided as aids in determining the optimal input capacitor solution.
Active Pulse Positioning Modulated PWM Operation
The ISL6323A uses a proprietary Active Pulse Positioning (APP) modulation scheme to control the internal PWM signals that command each channel's driver to turn their upper and lower MOSFETs on and off. The time interval in which a PWM signal can occur is generated by an internal clock, whose cycle time is the inverse of the switching frequency set by the resistor between the FS pin and ground. The advantage of Intersil's proprietary Active Pulse Positioning (APP) modulator is that the PWM signal has the ability to turn on at any point during this PWM time interval,
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inductor current, IL. This sensed current, ISEN, is simply a scaled version of the inductor current.
EXTERNAL CIRCUIT APA
ISL6323A INTERNAL CIRCUIT
CAPA
RAPA
VAPA,TRIP + COMP
100A
+ APA TO APA CIRCUITRY PWM SWITCHING PERIOD IL
FIGURE 3. ADAPTIVE PHASE ALIGNMENT DETECTION
+
-
The APA trip level is the amount of DC offset between the COMP pin and the APA pin. This is the voltage excursion that the APA and COMP pins must have during a transient event to activate the Adaptive Phase Alignment circuitry. This APA trip level is set through a resistor, RAPA, that connects from the APA pin to the COMP pin. A 100A current flows across RAPA into the APA pin to set the APA trip level as described in Equation 4. An APA trip level of 500mV is recommended for most applications. A 0.1F capacitor, CAPA, should also be placed across the RAPA resistor to help with noise immunity.
V APA, TRIP = R APA 100 x 10
-6
PWM Operation
The timing of each core channel is set by the number of active channels. Channel detection on the ISEN3- and ISEN4- pins selects 2-Channel to 4-Channel operation for the ISL6323A. The switching cycle is defined as the time between PWM pulse termination signals of each channel. The cycle time of the pulse signal is the inverse of the switching frequency set by the resistor between the FS pin and ground. The PWM signals command the MOSFET driver to turn on/off the channel MOSFETs. For 4-channel operation, the channel firing order is 1-2-3-4: PWM3 pulse happens 1/4 of a cycle after PWM4, PWM2 output follows another 1/4 of a cycle after PWM3, and PWM1 delays another 1/4 of a cycle after PWM2. For 3-channel operation, the channel firing order is 1-2-3. Connecting ISEN4- to VCC selects three channel operation and the pulse times are spaced in 1/3 cycle increments. If ISEN3- is connected to VCC, 2-Channel operation is selected and the PWM2 pulse happens 1/2 of a cycle after PWM1 pulse.
Continuous Current Sampling
In order to realize proper current-balance, the currents in each channel are sampled continuously every switching cycle. During this time, the current-sense amplifier uses the ISEN inputs to reproduce a signal proportional to the
-
LOW PASS FILTER
ERROR AMPLIFIER
ISEN
TIME
FIGURE 4. CONTINUOUS CURRENT SAMPLING
(EQ. 4)
The ISL6323A supports Inductor DCR current sensing to continuously sample each channel's current for channel-current balance. The internal circuitry, shown in Figure 3 represents Channel N of an N-Channel converter. This circuitry is repeated for each channel in the converter, but may not be active depending on how many channels are operating. Inductor windings have a characteristic distributed resistance or DCR (Direct Current Resistance). For simplicity, the inductor DCR is considered as a separate lumped quantity, as shown in Figure 5. The channel current ILn, flowing through the inductor, passes through the DCR. Equation 5 shows the S-domain equivalent voltage, VL, across the inductor.
V L ( s ) = I L ( s L + DCR ) n (EQ. 5)
A simple R-C network across the inductor (R1, R2 and C) extracts the DCR voltage, as shown in Figure 5. The voltage across the sense capacitor, VC, can be shown to be proportional to the channel current ILn, shown in Equation 6.
sL ------------- + 1 DCR V C ( s ) = ------------------------------------------------------- K DCR I L n ( R1 R2 ) s ----------------------- C + 1 R1 + R2 (EQ. 6)
Where:
R2 K = -------------------R2 + R1 (EQ. 7)
If the R-C network components are selected such that the RC time constant matches the inductor L/DCR time constant (see Equation 8), then VC is equal to the voltage drop across the DCR multiplied by the ratio of the resistor divider, K. If a resistor divider is not being used, the value for K is 1.
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VIN UGATE(n) MOSFET DRIVER LGATE(n) L DCR VOUT COUT I L
n
The North Bridge regulator samples the load current in the same manner as the Core regulator does. The RSET resistor will program all the effective internal RISEN resistors to the same value.
INDUCTOR C R2 RSET VL(s) + +
Channel-Current Balance
One important benefit of multi-phase operation is the thermal advantage gained by distributing the dissipated heat over multiple devices and greater area. By doing this the designer avoids the complexity of driving parallel MOSFETs and the expense of using expensive heat sinks and exotic magnetic materials. In order to realize the thermal advantage, it is important that each channel in a multi-phase converter be controlled to carry about the same amount of current at any load level. To achieve this, the currents through each channel must be sampled every switching cycle. The sampled currents, In, from each active channel are summed together and divided by the number of active channels. The resulting cycle average current, IAVG, provides a measure of the total load-current demand on the converter during each switching cycle. Channel-current balance is achieved by comparing the sampled current of each channel to the cycle average current, and making the proper adjustment to each channel pulse width based on the error. Intersil's patented current-balance method is illustrated in Figure 6, with error correction for Channel 1 represented. In the figure, the cycle average current, IAVG, is compared with the Channel 1 sample, I1, to create an error signal IER. The filtered error signal modifies the pulse width commanded by VCOMP to correct any unbalance and force IER toward zero. The same method for error signal correction is applied to each active channel.
VCOMP + + PWM1 TO GATE CONTROL LOGIC
VC(s)
R1
ISL6323A INTERNAL CIRCUIT In
SAMPLE + VC(s) RISEN ISEN +
ISENnISENn+ VCC RSET
CSET
FIGURE 5. INDUCTOR DCR CURRENT SENSING CONFIGURATION
.
R1 R2 L ------------- = -------------------- C R1 + R2 DCR
(EQ. 8)
The capacitor voltage VC, is then replicated across the effective internal sense resistor, RISEN. This develops a current through RISEN which is proportional to the inductor current. This current, ISEN, is continuously sensed and is then used by the controller for load-line regulation, channel-current balancing, and overcurrent detection and limiting. Equation 9 shows that the proportion between the channel current, IL, and the sensed current, ISEN, is driven by the value of the effective sense resistance, RISEN, and the DCR of the inductor.
DCR I SEN = I L ----------------R ISEN (EQ. 9)
FILTER f(s) IER
MODULATOR RAMP WAVEFORM
-
I4 IAVG
+
/N
I3 I2
The effective internal RISEN resistance is important to the current sensing process because it sets the gain of the load line regulation loop when droop is enabled as well as the gain of the channel-current balance loop and the overcurrent trip level. The effective internal RISEN resistance is user programmable and is set through use of the RSET pin. Placing a single resistor, RSET, from the RSET pin to the VCC pin programs the effective internal RISEN resistance according to Equation 10.
3 R ISEN = --------- R SET 400 (EQ. 10)
I1
NOTE: Channel 3 and 4 are optional. FIGURE 6. CHANNEL-1 PWM FUNCTION AND CURRENTBALANCE ADJUSTMENT
VID Interface
The ISL6323A supports hybrid power control of AMD processors which operate from either a 6-bit parallel VID interface (PVI) or a serial VID interface (SVI). The VID1/SEL pin is used to command the ISL6323A into either the PVI mode or the SVI mode. Whenever the EN pin is held LOW, both the
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multi-phase Core and single-phase North Bridge Regulators are disabled and the ISL6323A is continuously sampling voltage on the VID1/SEL pin. When the EN pin is toggled HIGH, the status of the VID1/SEL pin will latch the ISL6323A into either PVI or SVI mode. This latching occurs on the rising edge of the EN signal.If the VID1/SEL pin is held LOW during the latch, the ISL6323A will be placed into SVI mode. If the VID1/SEL pin is held HIGH during the latch, the ISL6323A will be placed into PVI mode. For the ISL6323A to properly enter into either mode, the level on the VID1/SEL pin must be stable no less than 1s prior to the EN signal transitioning from low to high.
TABLE 1. 6-BIT PARALLEL VID CODES (Continued) VID5 0 0 0 0 0 0 0 1 1 1 1 1 1 VREF 1.5500 1.5250 1.5000 1.4750 1.4500 1.4250 1.4000 1.3750 1.3500 1.3250 1.3000 1.2750 1.2500 1.2250 1.2000 1.1750 1.1500 1.1250 1.1000 1.0750 1.0500 1.0250 1.0000 0.9750 0.9500 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 VID4 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 VID3 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 VID2 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 VID1 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 VID0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 VREF 0.9250 0.9000 0.8750 0.8500 0.8250 0.8000 0.7750 0.7625 0.7500 0.7375 0.7250 0.7125 0.7000 0.6875 0.6750 0.6625 0.6500 0.6375 0.6250 0.6125 0.6000 0.5875 0.5750 0.5625 0.5500 0.5375 0.5250 0.5125 0.5000 0.4875 0.4750 0.4625 0.4500 0.4375 0.4250 0.4125 0.4000 0.3875 0.3750
6-Bit Parallel VID Interface (PVI)
With the ISL6323A in PVI mode, the single-phase North Bridge regulator is disabled. Only the multi-phase controller is active in PVI mode to support uniplane VDD only processors. Table 1 shows the 6-bit parallel VID codes and the corresponding reference voltage.
TABLE 1. 6-BIT PARALLEL VID CODES VID5 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 VID4 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 VID3 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 VID2 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 VID1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 VID0 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0
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Serial VID Interface (SVI)
The on-board Serial VID interface (SVI) circuitry allows the processor to directly drive the core voltage and Northbridge voltage reference level within the ISL6323A. The SVC and SVD states are decoded with direction from the PWROK and VFIXEN inputs as described in the sections that follow. The ISL6323A uses a digital to analog converter (DAC) to generate a reference voltage based on the decoded SVI value. See Figure 7 for a simple SVI interface timing diagram. PRE-PWROK METAL VID Typical motherboard start-up occurs with the VFIXEN input low. The controller decodes the SVC and SVD inputs to determine the Pre-PWROK metal VID setting. Once the POR circuitry is satisfied, the ISL6323A begins decoding the inputs per Table 2. Once the EN input exceeds the rising enable threshold, the ISL6323A saves the Pre-PWROK metal VID value in an on-board holding register and passes this target to the internal DAC circuitry.
TABLE 2. PRE-PWROK METAL VID CODES SVC 0 0 1 1 SVD 0 1 0 1 OUTPUT VOLTAGE (V) 1.1 1.0 0.9 0.8
The Pre-PWROK metal VID code is decoded and latched on the rising edge of the enable signal. Once enabled, the ISL6323A passes the Pre-PWROK metal VID code on to internal DAC circuitry. The internal DAC circuitry begins to ramp both the VDD and VDDNB planes to the decoded PrePWROK metal VID output level. The digital soft-start circuitry actually stair steps the internal reference to the target gradually over a fix interval. The controlled ramp of both output voltage planes reduces in-rush current during the soft-start interval. At the end of the soft-start interval, the VDDPWRGD output transitions high indicating both output planes are within regulation limits If the EN input falls below the enable falling threshold, the ISL6323A ramps the internal reference voltage down to near zero. The VDDPWRGD deasserts with the loss of enable. The VDD and VDDNB planes will linearly decrease to near zero.
TABLE 3. VFIXEN VID CODES SVC 0 0 1 1 SVD 0 1 0 1 OUTPUT VOLTAGE (V) 1.4 1.2 1.0 0.8
VFIX MODE In VFIX Mode, the SVC, SVD and VFIXEN inputs are fixed external to the controller through jumpers to either GND or VDDIO. These inputs are not expected to change, but the
1 VCC
2
3
4
5
6
7
8
9
10
11
12
SVC
SVD
ENABLE
PWROK METAL_VID VDD AND VDDNB V_SVI METAL_VID V_SVI
VDDPWRGD
VFIXEN
FIGURE 7. SVI INTERFACE TIMING DIAGRAM: TYPICAL PRE-PWROK METAL VID STAR-TUP
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ISL6323A is designed to support the potential change of state of these inputs. If VFIXEN is high, the IC decodes the SVC and SVD states per Table 3. Once enabled, the ISL6323A begins to soft-start both VDD and VDDNB planes to the programmed VFIX level. The internal soft-start circuitry slowly stair steps the reference up to the target value and this results in a controlled ramp of the power planes. Once soft-start has ended and both output planes are within regulation limits, the VDDPWRGD pin transitions high. If the EN input falls below the enable falling threshold, then the controller ramps both VDD and VDDNB down to near zero. SVI MODE Once the controller has successfully soft-started and VDDPWRGD transitions high, the Northbridge SVI interface can assert PWROK to signal the ISL6323A to prepare for SVI commands. The controller actively monitors the SVI interface for set VID commands to move the plane voltages to start-up VID values. Details of the SVI Bus protocol are provided in the AMD Design Guide for Voltage Regulator Controllers Accepting Serial VID Codes specification. Once the set VID command is received, the ISL6323A decodes the information to determine which plane and the VID target required (see Table 4). The internal DAC circuitry steps the required output plane voltage to the new VID level. During this time one or both of the planes could be targeted. In the event the core voltage plane, VDD, is commanded to power off by serial VID commands, the VDDPWRGD signal remains asserted. The Northbridge voltage plane must remain active during this time. If the PWROK input is deasserted, then the controller steps both VDD and VDDNB planes back to the stored PrePWROK metal VID level in the holding register from initial soft-start. No attempt is made to read the SVC and SVD inputs during this time. If PWROK is reasserted, then the on-board SVI interface waits for a set VID command. If VDDPWRGD deasserts during normal operation, both voltage planes are powered down in a controlled fashion. The internal DAC circuitry stair steps both outputs down to near zero.
TABLE 4. SERIAL VID CODES SVID[6:0] 000_0000b 000_0001b 000_0010b 000_0011b 000_0100b 000_0101b 000_0110b 000_0111b 000_1000b 000_1001b 000_1010b 000_1011b 000_1100b 000_1101b 000_1110b 000_1111b 001_0000b 001_0001b 001_0010b 001_0011b 001_0100b 001_0101b 001_0110b 001_0111b 001_1000b 001_1001b VOLTAGE (V) 1.5500 1.5375 1.5250 1.5125 1.5000 1.4875 1.4750 1.4625 1.4500 1.4375 1.4250 1.4125 1.4000 1.3875 1.3750 1.3625 1.3500 1.3375 1.3250 1.3125 1.3000 1.2875 1.2750 1.2625 1.2500 1.2375 SVID[6:0] 010_0000b 010_0001b 010_0010b 010_0011b 010_0100b 010_0101b 010_0110b 010_0111b 010_1000b 010_1001b 010_1010b 010_1011b 010_1100b 010_1101b 010_1110b 010_1111b 011_0000b 011_0001b 011_0010b 011_0011b 011_0100b 011_0101b 011_0110b 011_0111b 011_1000b 011_1001b VOLTAGE (V) 1.1500 1.1375 1.1250 1.1125 1.1000 1.0875 1.0750 1.0625 1.0500 1.0375 1.0250 1.0125 1.0000 0.9875 0.9750 0.9625 0.9500 0.9375 0.9250 0.9125 0.9000 0.8875 0.8750 0.8625 0.8500 0.8375 SVID[6:0] 100_0000b 100_0001b 100_0010b 100_0011b 100_0100b 100_0101b 100_0110b 100_0111b 100_1000b 100_1001b 100_1010b 100_1011b 100_1100b 100_1101b 100_1110b 100_1111b 101_0000b 101_0001b 101_0010b 101_0011b 101_0100b 101_0101b 101_0110b 101_0111b 101_1000b 101_1001b VOLTAGE (V) 0.7500 0.7375 0.7250 0.7125 0.7000 0.6875 0.6750 0.6625 0.6500 0.6375 0.6250 0.6125 0.6000 0.5875 0.5750 0.5625 0.5500 0.5375 0.5250 0.5125 0.5000 0.4875* 0.4750* 0.4625* 0.4500* 0.4375* SVID[6:0] 110_0000b 110_0001b 110_0010b 110_0011b 110_0100b 110_0101b 110_0110b 110_0111b 110_1000b 110_1001b 110_1010b 110_1011b 110_1100b 110_1101b 110_1110b 110_1111b 111_0000b 111_0001b 111_0010b 111_0011b 111_0100b 111_0101b 111_0110b 111_0111b 111_1000b 111_1001b VOLTAGE (V) 0.3500* 0.3375* 0.3250* 0.3125* 0.3000* 0.2875* 0.2750* 0.2625* 0.2500* 0.2375* 0.2250* 0.2125* 0.2000* 0.1875* 0.1750* 0.1625* 0.1500* 0.1375* 0.1250* 0.1125* 0.1000* 0.0875* 0.0750* 0.0625* 0.0500* 0.0375*
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TABLE 4. SERIAL VID CODES (Continued) SVID[6:0] 001_1010b 001_1011b 001_1100b 001_1101b 001_1110b 001_1111b VOLTAGE (V) 1.2250 1.2125 1.2000 1.1875 1.1750 1.1625 SVID[6:0] 011_1010b 011_1011b 011_1100b 011_1101b 011_1110b 011_1111b VOLTAGE (V) 0.8250 0.8125 0.8000 0.7875 0.7750 0.7625 SVID[6:0] 101_1010b 101_1011b 101_1100b 101_1101b 101_1110b 101_1111b VOLTAGE (V) 0.4250* 0.4125* 0.4000* 0.3875* 0.3750* 0.3625* SVID[6:0] 111_1010b 111_1011b 111_1100b 111_1101b 111_1110b 111_1111b VOLTAGE (V) 0.0250* 0.0125* OFF OFF OFF OFF
NOTE: *Indicates a VID not required for AMD Family 10h processors.
POWER SAVINGS MODE: PSI_L Bit 7 of the Serial VID codes transmitted as part of the 8-bit data phase over the SVI bus is allocated for the PSI_L. If Bit 7 is 0, then the processor is at an optimal load for the regulator to enter power savings mode. If Bit 7 is 1, then the regulator should not be in power savings mode. With the ISL6323A, Power Savings mode is realized through phase shedding. Once a Serial VID command with Bit 7 set to 0 is received, the ISL6323A will shed all phases in a sequential manner until only Channel 1 is switching. If active, Channel 4 will be shed first, followed by Channel 3 with Channel 2 being shed last. When a phase is shed, that phase will not go into a tri-state mode until that phase would have had its PWM go HIGH. When leaving Power Savings Mode, through the reception of a Serial VID command with Bit 7 set to 1, the ISL6323A will sequentially turn on phases starting with Phase 2. When a phase is being reactivated, it will not leave a tri-state until the PWM of that phase goes HIGH. If, while in Power Savings Mode, a Serial VID command is received that forces a VID level change while maintaining Bit 7 at 0, the ISL6323A will first exit the Power Savings Mode state as previously described. The output voltage will then be stepped up or down to the appropriate VID level. Finally, the ISL6323A will then re-enter Power Savings Mode.
.
V OUT = V REF - V OFS - V DROOP
(EQ. 11)
The ISL6323A incorporates differential remote-sense amplification in the feedback path. The differential sensing removes the voltage error encountered when measuring the output voltage relative to the controller ground reference point resulting in a more accurate means of sensing output voltage.
EXTERNAL CIRCUIT FS RFS COMP DROOP CONTROL TO OSCILLATOR ISL6323A INTERNAL CIRCUIT
CC 8 IAVG RC FB IOFS + + (VDROOP + VOFS) VCOMP ERROR AMPLIFIER
RFB
2k
VSEN + VOUT -
VID DAC
Voltage Regulation
The integrating compensation network shown in Figure 8 insures that the steady-state error in the output voltage is limited only to the error in the reference voltage and offset errors in the OFS current source, remote-sense and error amplifiers. Intersil specifies the guaranteed tolerance of the ISL6323A to include the combined tolerances of each of these elements. The output of the error amplifier, VCOMP, is used by the modulator to generate the PWM signals. The PWM signals control the timing of the Internal MOSFET drivers and regulate the converter output so that the voltage at FB is equal to the voltage at REF. This will regulate the output voltage to be equal to Equation 11. The internal and external circuitry that controls voltage regulation is illustrated in Figure 8. 18
RGND
FIGURE 8. OUTPUT VOLTAGE AND LOAD-LINE REGULATION WITH OFFSET ADJUSTMENT
Load-Line (Droop) Regulation
By adding a well controlled output impedance, the output voltage can effectively be level shifted in a direction which works to achieve a cost-effective solution can help to reduce the output-voltage spike that results from fast load-current demand changes. The magnitude of the spike is dictated by the ESR and ESL of the output capacitors selected. By positioning the no-load voltage level near the upper specification limit, a larger negative spike can be sustained without crossing the lower limit. By adding a well controlled output impedance, the
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output voltage under load can effectively be level shifted down so that a larger positive spike can be sustained without crossing the upper specification limit. As shown in Figure 8, with the FS resistor tied ground, a current eight times the average current of all active channels, 8*IAVG, flows from FB through a load-line regulation resistor RFB. The resulting voltage drop across RFB is proportional to the output current, effectively creating an output voltage droop with a steady-state value defined as in Equation 12:
V DROOP = I AVG R FB (EQ. 12)
VDIFF VOFS +
RFB
VREF E/A
FB IOFS
VCC + OFS ISL6323A GND 0.3V +
The regulated output voltage is reduced by the droop voltage VDROOP. The output voltage as a function of load current is shown in Equation 13.
I OUT 400 1 V OUT = V REF - V OFS - ------------- DCR --------- -------------- K R FB 3 R N SET (EQ. 13)
ROFS
1.6V
VCC
In Equation 13, VREF is the reference voltage, VOFS is the programmed offset voltage, IOUT is the total output current of the converter, KI is an internal gain determined by the RSET resistor connected to the RSET pin (KI is defined in Equation 10), K is the DC gain of the RC filter across the inductor (K is defined in Equation 7), N is the number of active channels, and DCR is the Inductor DCR value.
FIGURE 9. NEGATIVE OFFSET OUTPUT VOLTAGE PROGRAMMING
VOUT + VOFS -
Output-Voltage Offset Programming
The ISL6323A allows the designer to accurately adjust the offset voltage by connecting a resistor, ROFS, from the OFS pin to VCC or GND. When ROFS is connected between OFS and VCC, the voltage across it is regulated to 1.6V. This causes a proportional current (IOFS) to flow into the FB pin and out of the OFS pin. If ROFS is connected to ground, the voltage across it is regulated to 0.3V, and IOFS flows into the OFS pin and out of the FB pin. The offset current flowing through the resistor between VDIFF and FB will generate the desired offset voltage which is equal to the product (IOFS x RFB). These functions are shown in Figures 9 and 10.
RFB
VREF E/A
FB IOFS
+ OFS ROFS ISL6323A GND GND 0.3V +
1.6V
Once the desired output offset voltage has been determined, use the formulas in Equations 14 and 15 to set ROFS: For Positive Offset (connect ROFS to GND):
0.3 x R FB R OFS = -------------------------V OFFSET (EQ. 14)
VCC
FIGURE 10. POSITIVE OFFSET OUTPUT VOLTAGE PROGRAMMING
Dynamic VID
The AMD processor does not step the output voltage commands up or down to the target voltage, but instead passes only the target voltage to the ISL6323A through either the PVI or SVI interface. The ISL6323A manages the resulting VID-on-the-Fly transition in a controlled manner, supervising a safe output voltage transition without discontinuity or disruption. The ISL6323A begins slewing the DAC at 3.25mV/s until the DAC and target voltage are
For Negative Offset (connect ROFS to VCC):
1.6 x R FB R OFS = -------------------------V OFFSET (EQ. 15)
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equal. Thus, the total time required for a dynamic VID transition is dependent only on the size of the DAC change. To further improve dynamic VID performance, ISL6323A also implements a proprietary DAC smoothing feature. The external series RC components connected between DVC and FB limit any stair-stepping of the output voltage during a VID-on-the-Fly transition. the error amplifier R-C components between the FB and COMP pins.
V IN K1 = ---------V PP K1 A = ---------------K1 - 1 (EQ. 16)
R RCOMP = A x R C CC C RCOMP = ------A
(EQ. 17) (EQ. 18)
Compensating Dynamic VID Transitions
During a VID transition, the resulting change in voltage on the FB pin and the COMP pin causes an AC current to flow through the error amplifier compensation components from the FB to the COMP pin. This current then flows through the feedback resistor, RFB, and can cause the output voltage to overshoot or undershoot at the end of the VID transition. In order to ensure the smooth transition of the output voltage during a VID change, a VID-on-the-fly compensation network is required. This network is composed of a resistor and capacitor in series, RDVC and CDVC, between the DVC and the FB pin.
RFB VSEN IDVC CC CDVC DVC RDVC FB COMP IC RC
Advanced Adaptive Zero Shoot-Through Deadtime Control (Patent Pending)
The integrated drivers incorporate a unique adaptive deadtime control technique to minimize deadtime, resulting in high efficiency from the reduced freewheeling time of the lower MOSFET body-diode conduction, and to prevent the upper and lower MOSFETs from conducting simultaneously. This is accomplished by ensuring either rising gate turns on its MOSFET with minimum and sufficient delay after the other has turned off. During turn-off of the lower MOSFET, the PHASE voltage is monitored until it reaches a -0.3V/+0.8V (forward/reverse inductor current). At this time the UGATE is released to rise. An auto-zero comparator is used to correct the rDS(ON) drop in the phase voltage preventing false detection of the -0.3V phase level during rDS(ON) conduction period. In the case of zero current, the UGATE is released after 35ns delay of the LGATE dropping below 0.5V. When LGATE first begins to transition low, this quick transition can disturb the PHASE node and cause a false trip, so there is 20ns of blanking time once LGATE falls until PHASE is monitored. Once the PHASE is high, the advanced adaptive shoot-through circuitry monitors the PHASE and UGATE voltages during a PWM falling edge and the subsequent UGATE turn-off. If either the UGATE falls to less than 1.75V above the PHASE or the PHASE falls to less than +0.8V, the LGATE is released to turn-on.
IDVC = IC
+
VDAC+RGND ERROR AMPLIFIER
ISL6323A INTERNAL CIRCUIT
FIGURE 11. DYNAMIC VID COMPENSATION NETWORK
This VID-on-the-fly compensation network works by sourcing AC current into the FB node to offset the effects of the AC current flowing from the FB to the COMP pin during a VID transition. To create this compensation current the ISL6323A sets the voltage on the DVC pin to be 2x the voltage on the REF pin. Since the error amplifier forces the voltage on the FB pin and the REF pin to be equal, the resulting voltage across the series RC between DVC and FB is equal to the REF pin voltage. The RC compensation components, RDVC and CDVC, can then be selected to create the desired amount of compensation current. The amount of compensation current required is dependant on the modulator gain of the system, K1, and the error amplifier R-C components, RC and CC, that are in series between the FB and COMP pins. Use Equations 17, 18 and 19 to calculate the RC component values, RDVC and CDVC, for the VID-on-the-fly compensation network. For these equations: VIN is the input voltage for the power train; VP-P is the oscillator ramp amplitude (1.5V); and RC and CC are 20
Initialization
Prior to initialization, proper conditions must exist on the EN, VCC, PVCC1_2, PVCC_NB, ISEN3-, and ISEN4- pins. When the conditions are met, the controller begins soft-start. Once the output voltage is within the proper window of operation, the controller asserts PGOOD.
Power-On Reset
The ISL6323A requires VCC, PVCC1_2, and PVCC_NB inputs to exceed their rising POR thresholds before the ISL6323A has sufficient bias to guarantee proper operation. The bias voltage applied to VCC must reach the internal power-on reset (POR) rising threshold. Once this threshold is reached, the ISL6323A has enough bias to begin checking the driver POR inputs, EN, and channel detect portions of the initialization cycle. Hysteresis between the rising and falling thresholds assure the ISL6323A will not advertently
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ISL6323A INTERNAL CIRCUIT EXTERNAL CIRCUIT VCC PVCC1_2
If the controller is configured for 2-phase CORE operation, then the resistor divider can be used for sequencing the controller with another voltage rail. The resistor divider to EN should be selected using a similar approach as the previous driver discussion. The EN pin is also used to force the ISL6323A into either PVI or SVI mode. The mode is set upon the rising edge of the EN signal. When the voltage on the EN pin rises above 0.86V, the mode will be set depending upon the status of the VID1/SEL pin.
PVCC_NB +12V POR CIRCUIT
ENABLE COMPARATOR + EN
10.7k
Phase Detection
1.00k
The ISEN3- and ISEN4- pins are monitored prior to soft-start to determine the number of active CORE channel phases. If ISEN4- is tied to VCC, the controller will configure the channel firing order and timing for 3-phase operation. If ISEN3- and ISEN4- are tied to VCC, the controller will set the channel firing order and timing for 2-phase operation (see "PWM Operation" on page 13 for details).
0.86V
ISEN3SOFT-START AND FAULT LOGIC CHANNEL DETECT ISEN4-
Soft-Start Output Voltage Targets
Once the POR and Phase Detect blocks and enable comparator are satisfied, the controller will begin the softstart sequence and will ramp the CORE and NB output voltages up to the SVI interface designated target level if the controller is set SVI mode. If set to PVI mode, the North Bridge regulator is disabled and the core is soft started to the level designated by the parallel VID code.
FIGURE 12. POWER SEQUENCING USING THRESHOLDSENSITIVE ENABLE (EN) FUNCTION
turn off unless the bias voltage drops substantially (see Electrical Specifications on page 6). The bias voltage applied to the PVCC1_2 and PVCC_NB pins power the internal MOSFET drivers of each output channel. In order for the ISL6323A to begin operation, both PVCC inputs must exceed their POR rising threshold to guarantee proper operation of the internal drivers. Hysteresis between the rising and falling thresholds assure that once enabled, the ISL6323A will not inadvertently turn off unless the PVCC bias voltage drops substantially (see "Electrical Specifications" on page 6). Depending on the number of active CORE channels determined by the Phase Detect block, the external driver POR checking is supported by the Enable Comparator.
SVI Mode
Prior to soft-starting both CORE and NB outputs, the ISL6323A must check the state of the SVI interface inputs to determine the correct target voltages for both outputs. When the controller is enabled, the state of the VFIXEN, SVD and SVC inputs are checked and the target output voltages set for both CORE and NB outputs are set by the DAC (see "Serial VID Interface (SVI)" on page 16). These targets will only change if the EN signal is pulled low or after a POR reset of VCC.
Enable Comparator
The ISL6323A features a dual function enable input (EN) for enabling the controller and power sequencing between the controller and external drivers or another voltage rail. The enable comparator holds the ISL6323A in shutdown until the voltage at EN rises above 0.86V. The enable comparator has about 110mV of hysteresis to prevent bounce. It is important that the driver ICs reach their rising POR level before the ISL6323A becomes enabled. The schematic in Figure 12 demonstrates sequencing the ISL6323A with the ISL66xx family of Intersil MOSFET drivers, which require 12V bias. When selecting the value of the resistor divider the driver maximum rising POR threshold should be used for calculating the proper resistor values. This will prevent improper sequencing events from creating false trips during soft-start.
Soft-Start
The soft-start sequence is composed of three periods, as shown in Figure 13. At the beginning of soft-start, the DAC immediately obtains the output voltage targets for both outputs by decoding the state of the SVI or PVI inputs. A 100s fixed delay time, TDA, proceeds the output voltage rise. After this delay period the ISL6323A will begin ramping both CORE and NB output voltages to the programmed DAC level at a fixed rate of 3.25mV/s. The amount of time required to ramp the output voltage to the final DAC voltage is referred to as TDB, and can be calculated as shown in Equation 19.
V DAC TDB = -----------------------------3 3.25 x 10 (EQ. 19)
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After the DAC voltage reaches the final VID setting, PGOOD will be set to high.
Fault Monitoring and Protection
The ISL6323A actively monitors both CORE and NB output voltages and currents to detect fault conditions. Fault monitors trigger protective measures to prevent damage to either load. One common power good indicator is provided for linking to external system monitors. The schematic in Figure 15 outlines the interaction between the fault monitors and the power good signal.
VNB 400mV/DIV
VCORE 400mV/DIV
TDA EN 5V/DIV
TDB
Power Good Signal
The power good pin (VDDPWRGD) is an open-drain logic output that signals whether or not the ISL6323A is regulating both NB and CORE output voltages within the proper levels, and whether any fault conditions exist. This pin should be tied to a +5V source through a resistor. During shutdown and soft-start, VDDPWRGD pulls low and releases high after a successful soft-start and both output voltages are operating between the undervoltage and overvoltage limits. PGOOD transitions low when an undervoltage, overvoltage, or overcurrent condition is detected on either output or when the controller is disabled by a POR reset or EN. In the event of an overvoltage or overcurrent condition, the controller latches off and PGOOD will not return high. Pending a POR reset of the ISL6323A and successful soft-start, the PGOOD will return high.
VDDPWRGD 5V/DIV
100s/DIV
FIGURE 13. SOFT-START WAVEFORMS
Pre-Biased Soft-Start
The ISL6323A also has the ability to start up into a pre-charged output, without causing any unnecessary disturbance. The FB pin is monitored during soft-start, and should it be higher than the equivalent internal ramping reference voltage, the output drives hold both MOSFETs off. Once the internal ramping reference exceeds the FB pin potential, the output drives are enabled, allowing the output to ramp from the pre-charged level to the final level dictated by the DAC setting. Should the output be pre-charged to a level exceeding the DAC setting, the output drives are enabled at the end of the soft-start period, leading to an abrupt correction in the output voltage down to the DAC-set level. Both CORE and NB output support start-up into a pre-charged output.
OUTPUT PRECHARGED ABOVE DAC LEVEL
Overvoltage Protection
The ISL6323A constantly monitors the sensed output voltage on the VSEN pin to detect if an overvoltage event occurs. When the output voltage rises above the OVP trip level and exceeds the PGOOD OV limit actions are taken by the ISL6323A to protect the microprocessor load. At the inception of an overvoltage event, both on-board lower gate pins are commanded low as are the active PWM outputs to the external drivers, the PGOOD signal is driven low, and the ISL6323A latches off normal PWM action. This turns on the all of the lower MOSFETs and pulls the output voltage below a level that might cause damage to the load. The lower MOSFETs remain driven ON until VDIFF falls below 400mV. The ISL6323A will continue to protect the load in this fashion as long as the overvoltage condition recurs. Once an overvoltage condition ends the ISL6323A latches off, and must be reset by toggling POR, before a soft-start can be re-initiated.
OUTPUT PRECHARGED BELOW DAC LEVEL VCORE 400mV/DIV
EN 5V/DIV
100s/DIV
FIGURE 14. SOFT-START WAVEFORMS FOR ISL6323ABASED MULTIPHASE CONVERTER
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Open Sense Line Protection
OCL + 100A INB 142A I1
OCP +
REPEAT FOR EACH CORE CHANNEL
OCP +
100A IAVG CORE ONLY
In the case that either of the remote sense lines, VSEN or GND, become open, the ISL6323A is designed to detect this and shut down the controller. This event is detected by monitoring small currents that are fed out the VSEN and RGND pins. In the event of an open sense line fault, the controller will continue to remain off until the fault goes away, at which point the controller will re-initiate a soft-start sequence.
NB ONLY SOFT-START, FAULT AND CONTROL LOGIC NB ONLY 1.8V
Overcurrent Protection
The ISL6323A takes advantage of the proportionality between the load current and the average current, IAVG, to detect an overcurrent condition. See "Continuous Current Sampling" on page 13 and "Channel-Current Balance" on page 14 for more detail on how the average current is measured. Once the average current exceeds 100A, a comparator triggers the converter to begin overcurrent protection procedures. The Core regulator and the North Bridge regulator have the same type of overcurrent protection.
+ OVP
ISEN_NB+ DAC - 300mV CORE ONLY 1.8V + OVP
UV +
DAC + 250mV
VDDPWRGD
OV +
The overcurrent trip threshold is dictated by the DCR of the inductors, the number of active channels, the DC gain of the inductor RC filter and the RSET resistor. The overcurrent trip threshold is shown in Equation 20.
V IN - N V OUT V OUT N 1 3 I OCP = 100A ------------- --- --------- R SET - ---------------------------------------- -------------- 400 V IN 2 L fS DCR K (EQ. 20)
VSEN DAC - 300mV ISL6323A INTERNAL CIRCUITRY UV +
Where:
R2 K = -------------------R1 + R2
FIGURE 15. POWER GOOD AND PROTECTION CIRCUITRY
See "Continuous Current Sampling" on page 13.
Pre-POR Overvoltage Protection
Prior to PVCC and VCC exceeding their POR levels, the ISL6323A is designed to protect either load from any overvoltage events that may occur. This is accomplished by means of an internal 10k resistor tied from PHASE to LGATE, which turns on the lower MOSFET to control the output voltage until the overvoltage event ceases or the input power supply cuts off. For complete protection, the low side MOSFET should have a gate threshold well below the maximum voltage rating of the load/microprocessor. In the event that during normal operation the PVCC or VCC voltage falls back below the POR threshold, the pre-POR overvoltage protection circuitry reactivates to protect from any more pre-POR overvoltage events.
fSW = Switching Frequency
Equation 20 is valid for both the Core regulator and the North Bridge regulator. This equation includes the DC load current as well as the total ripple current contributed by all the phases. For the North Bridge regulator, N is 1. During soft-start, the overcurrent trip point is boosted by a factor of 1.4. Instead of comparing the average measured current to 100A, the average current is compared to 140A. Immediately after soft-start is over, the comparison level changes to 100A. This is done to allow for start-up into an active load while still supplying output capacitor in-rush current. CORE REGULATOR OVERCURRENT At the beginning of overcurrent shutdown, the controller sets all of the UGATE and LGATE signals low, puts PWM3 and PWM4 (if active) in a high-impedance state, and forces VDDPWRGD low. This turns off all of the upper and lower MOSFETs. The system remains in this state for fixed period of 12ms. If the controller is still enabled at the end of this wait
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Undervoltage Detection
The undervoltage threshold is set at VDAC - 300mV typical. When the output voltage (VSEN-RGND) is below the undervoltage threshold, PGOOD gets pulled low. No other action is taken by the controller. PGOOD will return high if the output voltage rises above VDAC - 250mV typical.
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ISL6323A
period, it will attempt a soft-start, as shown in Figure 16. If the fault remains, the trip-retry cycles will continue until either the fault is cleared or for a total of seven attempts. If the fault is not cleared on the final attempt, the controller disables UGATE and LGATE signals for both Core and North Bridge and latches off requiring a POR of VCC to reset the ISL6323A. It is important to note that during soft start, the overcurrent trip point is increased by a factor of 1.4. If the fault draws enough current to trip overcurrent during normal run mode, it may not draw enough current during the soft start ramp period to trip overcurrent while the output is ramping up. If a fault of this type is affecting the output, then the regulator will complete soft start and the trip-retry counter will be reset to zero. Once the regulator has completed soft start, the overcurrent trip point will return to it's nominal setting and an overcurrent shutdown will be initiated. This will result in a continuous hiccup mode. Note that the energy delivered during trip-retry cycling is much less than during full-load operation, so there is no thermal hazard. OVERCURRENT PROTECTION IN POWER SAVINGS MODE While in Power Savings Mode, the OCP trip point will be lower than when running in Normal Mode. Equation 20, with N = 1, will yield the OCP trip point for the Core regulator while in Power Savings mode. If an overcurrent event should occur while the system is in Power Savings Mode, the ISL6323A will restart in the Normal state with the PSI_L bit set to 1.
Individual Channel Overcurrent Limiting
The ISL6323A has the ability to limit the current in each individual channel of the Core regulator without shutting down the entire regulator. This is accomplished by continuously comparing the sensed currents of each channel with a constant 140A OCL reference current. If a channel's individual sensed current exceeds this OCL limit, the UGATE signal of that channel is immediately forced low, and the LGATE signal is forced high. This turns off the upper MOSFET(s), turns on the lower MOSFET(s), and stops the rise of current in that channel, forcing the current in the channel to decrease. That channel's UGATE signal will not be able to return high until the sensed channel current falls back below the 140A reference.
OUTPUT CURRENT, 50A/DIV
General Design Guide
0A
OUTPUT VOLTAGE, 500mV/DIV
This design guide is intended to provide a high-level explanation of the steps necessary to create a multiphase power converter. It is assumed that the reader is familiar with many of the basic skills and techniques referenced below. In addition to this guide, Intersil provides complete reference designs that include schematics, bills of materials, and example board layouts for all common microprocessor applications.
Power Stages
0V 3ms/DIV
FIGURE 16. OVERCURRENT BEHAVIOR IN HICCUP MODE
NORTH BRIDGE REGULATOR OVERCURRENT The overcurrent shutdown sequence for the North Bridge regulator is identical to the Core regulator with the exception that it is a single phase regulator and will only disable the MOSFET drivers for the North Bridge. Once 7 retry attempts have been executed unsuccessfully, the controller will disable UGATE and LGATE signals for both Core and North Bridge and will latch off requiring a POR of VCC to reset the ISL6323A. Note that the energy delivered during trip-retry cycling is much less than during full-load operation, so there is no thermal hazard.
The first step in designing a multi-phase converter is to determine the number of phases. This determination depends heavily on the cost analysis which in turn depends on system constraints that differ from one design to the next. Principally, the designer will be concerned with whether components can be mounted on both sides of the circuit board, whether through-hole components are permitted, the total board space available for power-supply circuitry, and the maximum amount of load current. Generally speaking, the most economical solutions are those in which each phase handles between 25A and 30A. All surface-mount designs will tend toward the lower end of this current range. If through-hole MOSFETs and inductors can be used, higher per-phase currents are possible. In cases where board space is the limiting constraint, current can be pushed as high as 40A per phase, but these designs require heat sinks and forced air to cool the MOSFETs, inductors and heat-dissipating surfaces.
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MOSFETS The choice of MOSFETs depends on the current each MOSFET will be required to conduct, the switching frequency, the capability of the MOSFETs to dissipate heat, and the availability and nature of heat sinking and air flow. LOWER MOSFET POWER CALCULATION The calculation for power loss in the lower MOSFET is simple, since virtually all of the loss in the lower MOSFET is due to current conducted through the channel resistance (rDS(ON)). In Equation 21, IM is the maximum continuous output current, IP-P is the peak-to-peak inductor current and d is the duty cycle (VOUT/VIN).
2 I L ( P-P ) ( 1 - d ) I M 2 P LOW, 1 = r DS ( ON ) ----- ( 1 - d ) + --------------------------------------12 N
At turn on, the upper MOSFET begins to conduct and this transition occurs over a time t2. In Equation 24, the approximate power loss is PUP,2.
I M I PP t 2 P UP, 2 V IN ----- - -------- ---- f S 2 2 N (EQ. 24)
A third component involves the lower MOSFET reverserecovery charge, Qrr. Since the inductor current has fully commutated to the upper MOSFET before the lower-MOSFET body diode can recover all of Qrr, it is conducted through the upper MOSFET across VIN. The power dissipated as a result is PUP,3.
P UP,3 = V IN Q rr f S (EQ. 25)
(EQ. 21)
An additional term can be added to the lower-MOSFET loss equation to account for additional loss accrued during the dead time when inductor current is flowing through the lower-MOSFET body diode. This term is dependent on the diode forward voltage at IM, VD(ON), the switching frequency, fS, and the length of dead times, td1 and td2, at the beginning and the end of the lower-MOSFET conduction interval respectively.
I I I (EQ. 22) M P LOW, 2 = V D ( ON ) f S ------ + I P-P t + M P-P t ----------d1 ------ - ----------- d2 2 N 2 N
Finally, the resistive part of the upper MOSFET is given in Equation 26 as PUP,4.
2 I M I PP P UP,4 r DS ( ON ) ----- d + --------12 N 2
(EQ. 26)
The total power dissipated by the upper MOSFET at full load can now be approximated as the summation of the results from Equations 23, 24, 25 and 26. Since the power equations depend on MOSFET parameters, choosing the correct MOSFETs can be an iterative process involving repetitive solutions to the loss equations for different MOSFETs and different switching frequencies.
The total maximum power dissipated in each lower MOSFET is approximated by the summation of PLOW,1 and PLOW,2. UPPER MOSFET POWER CALCULATION In addition to rDS(ON) losses, a large portion of the upper-MOSFET losses are due to currents conducted across the input voltage (VIN) during switching. Since a substantially higher portion of the upper-MOSFET losses are dependent on switching frequency, the power calculation is more complex. Upper MOSFET losses can be divided into separate components involving the upper-MOSFET switching times, the lower-MOSFET body-diode reverse-recovery charge, Qrr, and the upper MOSFET rDS(ON) conduction loss. When the upper MOSFET turns off, the lower MOSFET does not conduct any portion of the inductor current until the voltage at the phase node falls below ground. Once the lower MOSFET begins conducting, the current in the upper MOSFET falls to zero as the current in the lower MOSFET ramps up to assume the full inductor current. In Equation 23, the required time for this commutation is t1 and the approximated associated power loss is PUP,1.
I M I P-P t 1 P UP,1 V IN ----- + --------- ---- f S N2 2 (EQ. 23)
Internal Bootstrap Device
All three integrated drivers feature an internal bootstrap schottky diode. Simply adding an external capacitor across the BOOT and PHASE pins completes the bootstrap circuit. The bootstrap function is also designed to prevent the bootstrap capacitor from overcharging due to the large negative swing at the PHASE node. This reduces voltage stress on the boot to phase pins. The bootstrap capacitor must have a maximum voltage rating above PVCC + 4V and its capacitance value can be chosen from Equation 27:
Q GATE C BOOT_CAP ------------------------------------V BOOT_CAP Q G1 * PVCC Q GATE = ----------------------------------- * N Q1 V GS1
(EQ. 27)
where QG1 is the amount of gate charge per upper MOSFET at VGS1 gate-source voltage and NQ1 is the number of control MOSFETs. The VBOOT_CAP term is defined as the allowable droop in the rail of the upper gate drive.
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1.6 1.4 1.2 CBOOT_CAP (F) 1. 0.8 0.6 QGATE = 100nC 0.4 50nC 0.2 20nC 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
P Qg_TOT = P Qg_Q1 + P Qg_Q2 + I Q VCC 3 P Qg_Q1 = -- Q G1 PVCC F SW N Q1 N PHASE 2 P Qg_Q2 = Q G2 PVCC F SW N Q2 N PHASE
(EQ. 28)
3 I DR = -- Q G1 N + Q G2 N Q2 N PHASE F SW + I Q 2 Q1
(EQ. 29)
0.0 0.0
VBOOT_CAP (V)
FIGURE 17. BOOTSTRAP CAPACITANCE vs BOOT RIPPLE VOLTAGE
Gate Drive Voltage Versatility
The ISL6323A provides the user flexibility in choosing the gate drive voltage for efficiency optimization. The controller ties the upper and lower drive rails together. Simply applying a voltage from 5V up to 12V on PVCC sets both gate drive rail voltages simultaneously.
In Equations 28 and 29, PQg_Q1 is the total upper gate drive power loss and PQg_Q2 is the total lower gate drive power loss; the gate charge (QG1 and QG2) is defined at the particular gate to source drive voltage PVCC in the corresponding MOSFET data sheet; IQ is the driver total quiescent current with no load at both drive outputs; NQ1 and NQ2 are the number of upper and lower MOSFETs per phase, respectively; NPHASE is the number of active phases. The IQ*VCC product is the quiescent power of the controller without capacitive load and is typically 75mW at 300kHz.
PVCC BOOT D CGD RHI1 RLO1 UGATE G RG1 RGI1 CGS S PHASE Q1 CDS
Package Power Dissipation
When choosing MOSFETs it is important to consider the amount of power being dissipated in the integrated drivers located in the controller. Since there are a total of three drivers in the controller package, the total power dissipated by all three drivers must be less than the maximum allowable power dissipation for the QFN package. Calculating the power dissipation in the drivers for a desired application is critical to ensure safe operation. Exceeding the maximum allowable power dissipation level will push the IC beyond the maximum recommended operating junction temperature of +125C. The maximum allowable IC power dissipation for the 7x7 QFN package is approximately 3.5W at room temperature. See "Layout Considerations" on page 32 for thermal transfer improvement suggestions. When designing the ISL6323A into an application, it is recommended that the following calculation is used to ensure safe operation at the desired frequency for the selected MOSFETs. The total gate drive power losses, PQg_TOT, due to the gate charge of MOSFETs and the integrated driver's internal circuitry and their corresponding average driver current can be estimated with Equations 28 and 29, respectively.
FIGURE 18. TYPICAL UPPER-GATE DRIVE TURN-ON PATH
PVCC D CGD RHI2 RLO2 LGATE G RG2 RGI2 CGS S Q2 CDS
FIGURE 19. TYPICAL LOWER-GATE DRIVE TURN-ON PATH
The total gate drive power losses are dissipated among the resistive components along the transition path and in the bootstrap diode. The portion of the total power dissipated in the controller itself is the power dissipated in the upper drive path resistance (PDR_UP), the lower drive path resistance (PDR_UP), and in the boot strap diode (PBOOT). The rest of the power will be dissipated by the external gate resistors (RG1 and RG2) and the internal gate resistors (RGI1 and RGI2) of the MOSFETs. Figures 18 and 19 show the typical upper and lower gate drives turn-on transition path. The total
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power dissipation in the controller itself, PDR, can be roughly estimated as Equation 30:
P DR = P DR_UP + P DR_LOW + P BOOT + ( I Q VCC ) P Qg_Q1 P BOOT = --------------------3 R HI1 R LO1 P Qg_Q1 P DR_UP = -------------------------------------- + --------------------------------------- --------------------R HI1 + R EXT1 R LO1 + R EXT1 3 R LO2 R HI2 P Qg_Q2 P DR_LOW = -------------------------------------- + --------------------------------------- --------------------2 R HI2 + R EXT2 R LO2 + R EXT2 R GI1 R EXT1 = R G1 + ------------N Q1 R GI2 R EXT2 = R G2 + ------------N Q2 (EQ. 30)
3. Calculate the value for the RSET resistor using Equation 33: (Derived from Equation 20).
V IN - V NB V NB 400 DCR NB K R SET = --------- ----------------------------- I OCP + --------------------------------- ----------- 100A 2 L NB f SW V IN 3 NB
Where: K = 1
(EQ. 33)
4. Using Equation 34 (also derived from Equation 20), calculate the value of K for the Core regulator.
N 100A 3K = --------- R SET ----------------------------- ----------------------------------------------------------------------------------------------------------V IN - N V CORE V CORE DCR CORE 400 I OCP + -------------------------------------------- ------------------V IN CORE 2 L CORE f SW (EQ. 34)
5. Choose a capacitor value for the Core RC filters. A 0.1F capacitor is a recommended starting point. 6. Calculate the values for R1 and R2 for Core. Equations 35 and 36 will allow for their computation.
R2 Core K = ---------------------------------------------R1 + R2
Core Core
Inductor DCR Current Sensing Component Selection and RSET Value Calculation
With the single RSET resistor setting the value of the effective internal sense resistors for both the North Bridge and Core regulators, it is important to set the RSET value and the inductor RC filter gain, K, properly. See "Continuous Current Sampling" on page 13 and "Channel-Current Balance" on page 14 for more details on the application of the RSET resistor and the RC filter gain. There are 3 separate cases to consider when calculating these component values. If the system under design will never utilize the North Bridge regulator and the ISL6323 will always be in parallel mode, then follow the instructions for Case 3 and only calculate values for Core regulator components. For all three cases, use the expected VID voltage that would be used at TDC for Core and North Bridge for the VCORE and VNB variables, respectively. CASE 1
I NB I Core MAX DCR NB < -------------------------- DCR Core N MAX (EQ. 31)
(EQ. 35)
R1 R2 L Core Core Core ------------------------- = ---------------------------------------------- C Core DCR Core + R2 R1
Core Core
(EQ. 36)
CASE 2
I NB I Core MAX DCR NB > -------------------------- DCR Core N MAX (EQ. 37)
In Case 2, the DC voltage across the North Bridge inductor at full load is greater than the DC voltage across a single phase of the Core regulator while at full load. Here, the DC voltage across the North Bridge inductor must be scaled down to match the DC voltage across the Core inductors, which will be impressed across the ISEN pins without any gain. So, the R2 resistor for the Core inductor RC filters is left unpopulated and K = 1. 1. Choose a capacitor value for the Core RC filter. A 0.1F capacitor is a recommended starting point. 2. Calculate the value for resistor R1:
R1 L Core = ----------------------------------------------DCR Core C Core Core (EQ. 38)
In Case 1, the DC voltage across the North Bridge inductor at full load is less than the DC voltage across a single phase of the Core regulator while at full load. Here, the DC voltage across the Core inductors must be scaled down to match the DC voltage across the North Bridge inductor, which will be impressed across the ISEN_NB pins without any gain. So, the R2 resistor for the North Bridge inductor RC filter is left unpopulated and K = 1. 1. Choose a capacitor value for the North Bridge RC filter. A 0.1F capacitor is a recommended starting point. 2. Calculate the value for resistor R1 using Equation 32:
R1
NB
3. Calculate the value for the RSET resistor using Equation 39 (Derived from Equation 20).:
V IN - V CORE V CORE 400 DCR CORE K R SET = --------- -------------------------------------- I OCP + ------------------------------------------ ------------------- 100A V IN 3 CORE 2 L CORE f SW
Where: K = 1
(EQ. 39)
4. Using Equation 40 (also derived from Equation 20), calculate the value of K for the North bridge regulator.
100A 1 3 K = --------- R SET --------------------- -----------------------------------------------------------------------------V IN - V NB V NB DCR NB 400 I OCP + --------------------------------- ----------NB 2 L NB f SW V IN (EQ. 40)
L NB = ------------------------------------DCR NB C NB
(EQ. 32)
27
FN6878.0
ISL6323A
5. Choose a capacitor value for the North Bridge RC filter. A 0.1F capacitor is a recommended starting point. 6. Calculate the values for R1 and R2 for North Bridge. Equations 41 and 42 will allow for their computation.
R2 NB K = -----------------------------------R1 + R2
NB
(EQ. 41)
NB
R1 R2 L NB NB NB --------------------- = ------------------------------------ C NB DCR NB + R2 R1 NB NB
(EQ. 42)
NOTE: The values of RSET must be greater than 20k and less than 80k. For all of the 3 cases, if the calculated value of RSET is less than 20k, then either the OCP trip point needs to be increased or the inductor must be changed to an inductor with higher DCR. If the RSET resistor is greater than 80k, then a value of RSET that is less than 80k must be chosen and a resistor divider across both North Bridge and Core inductors must be set up with proper gain. This gain will represent the variable "K" in all equations. It is also very important that the RSET resistor be tied between the RSET pin and the VCC pin of the ISL6323.
CASE 3
I NB I Core MAX DCR NB = -------------------------- DCR Core N MAX (EQ. 43)
Inductor DCR Current Sensing Component Fine Tuning
VIN UGATE(n) MOSFET DRIVER LGATE(n) L DCR VOUT COUT I L n
For this Case, it is recommended that the overcurrent trip point for the North Bridge regulator be equal to the overcurrent trip point for the Core regulator divided by the number of core phases. 1. Choose a capacitor value for the North Bridge RC filter. A 0.1F capacitor is a recommended starting point. 2. Calculate the value for the North Bridge resistor R1:
R1
NB
R1
C R2
ISL6323A INTERNAL CIRCUIT
In KI 40kK I = ---------------R SET
L NB = ------------------------------------DCR NB C NB
(EQ. 44)
SAMPLE + VC(s) RISEN 2.4k +
3. Choose a capacitor value for the Core RC filter. A 0.1F capacitor is a recommended starting point. 5. Calculate the value for the Core resistor R1:
R1 L Core = ----------------------------------------------DCR Core C Core Core (EQ. 45)
ISEN
ISENnISENn+
RSET VCC RSET
6. Calculate the value for the RSET resistor using Equation 46:
V IN - V CORE V CORE 400 DCR CORE K R SET = --------- -------------------------------------- I OCP + ------------------------------------------ ------------------- V IN 100A 3 CORE 2 L CORE f SW
FIGURE 20. DCR SENSING CONFIGURATION
Where: K = 1
(EQ. 46)
7. Calculate the OCP trip point for the North Bridge regulator using Equation 47. If the OCP trip point is higher than desired, then the component values must be recalculated utilizing Case 1. If the OCP trip point is lower than desired, then the component values must be recalculated utilized Case 2.
V IN - V NB V NB 3 1 = 100A --------------------- --------- R SET + --------------------------------- ----------I OCP 2L DCR NB 400 NB NB f SW V IN (EQ. 47)
Due to errors in the inductance and/or DCR it may be necessary to adjust the value of R1 and R2 to match the time constants correctly. The effects of time constant mismatch can be seen in the form of droop overshoot or undershoot during the initial load transient spike, as shown in Figure 21. Follow the steps below to ensure the R-C and inductor L/DCR time constants are matched accurately. 1. If the regulator is not utilizing droop, modify the circuit by placing the frequency set resistor between FS and Ground for the duration of this procedure.
28
-
VC(s)
+
FN6878.0
In Case 3, the DC voltage across the North Bridge inductor at full load is equal to the DC voltage across a single phase of the Core regulator while at full load. Here, the full scale DC inductor voltages for both North Bridge and Core will be impressed across the ISEN pins without any gain. So, the R2 resistors for the Core and North Bridge inductor RC filters are left unpopulated and K = 1 for both regulators.
INDUCTOR VL(s) +
ISL6323A
2. Capture a transient event with the oscilloscope set to about L/DCR/2 (sec/div). For example, with L = 1H and DCR = 1m, set the oscilloscope to 500s/div. 3. Record V1 and V2 as shown in Figure 21.
RC
function, the gain of the current signal, and the value of the compensation components, RC and CC.
C2 (OPTIONAL)
CC
COMP
V2 V1 VOUT RFB
FB
ISL6323A
VSEN
ITRAN I
FIGURE 22. COMPENSATION CONFIGURATION FOR LOAD-LINE REGULATED ISL6323A CIRCUIT
FIGURE 21. TIME CONSTANT MISMATCH BEHAVIOR
4. Select new values, R1,NEW and R2,NEW, for the time constant resistors based on the original values, R1,OLD and R2,OLD, using Equations 48 and 49. V 1 (EQ. 48) R 1, NEW = R 1, OLD ---------V 2 V 1 R 2, NEW = R 2, OLD ---------V 2
(EQ. 49)
Since the system poles and zero are affected by the values of the components that are meant to compensate them, the solution to the system equation becomes fairly complicated. Fortunately, there is a simple approximation that comes very close to an optimal solution. Treating the system as though it were a voltage-mode regulator, by compensating the L-C poles and the ESR zero of the voltage mode approximation, yields a solution that is always stable with very close to ideal transient performance. Select a target bandwidth for the compensated system, f0. The target bandwidth must be large enough to assure adequate transient performance, but smaller than 1/3 of the per-channel switching frequency. The values of the compensation components depend on the relationships of f0 to the L-C pole frequency and the ESR zero frequency. For each of the following three, there is a separate set of equations for the compensation components. In Equation 51, L is the per-channel filter inductance divided by the number of active channels; C is the sum total of all output capacitors; ESR is the equivalent series resistance of the bulk output filter capacitance; and VP-P is the peak-to-peak sawtooth signal amplitude as described in the Electrical Specifications on page 6. Once selected, the compensation values in Equation 51 assure a stable converter with reasonable transient performance. In most cases, transient performance can be improved by making adjustments to RC. Slowly increase the value of RC while observing the transient performance on an oscilloscope until no further improvement is noted. Normally, CC will not need adjustment. Keep the value of CC from Equation 51 unless some performance issue is noted. The optional capacitor C2, is sometimes needed to bypass noise away from the PWM comparator (see Figure 22). Keep a position available for C2, and be prepared to install a highfrequency capacitor of between 22pF and 150pF in case any leading edge jitter problem is noted.
5. Replace R1 and R2 with the new values and check to see that the error is corrected. Repeat the procedure if necessary.
Loadline Regulation Resistor
The loadline regulation resistor, labeled RFB in Figure 8, sets the desired loadline required for the application. Equation 50 can be used to calculate RFB.
V DROOP MAX R FB = --------------------------------------------------------------------I OUT 400 MAX DCR --------- ------------------------- -------------- K N R SET 3 (EQ. 50)
Where RISEN is the 2.4k internal current sense resistor, KI is defined in Equation 10 and K is defined in Equation 7. If no loadline regulation is required, FS resistor should be tied between the FS pin and VCC. To choose the value for RFB in this situation, please refer to "Compensation Without Loadline Regulation" on page 30.
Compensation With Loadline Regulation
The load-line regulated converter behaves in a similar manner to a peak current mode controller because the two poles at the output filter L-C resonant frequency split with the introduction of current information into the control loop. The final location of these poles is determined by the system
29
FN6878.0
ISL6323A
.
Case 1:
1 ------------------------------- > f 0 2 LC 2 f 0 V P-P L C R C = R FB ---------------------------------------------------------0.66 V
IN
C ESR R 1 = R FB ------------------------------------------L C - C ESR L C - C ESR C 1 = ------------------------------------------R FB 0.75 V IN C 2 = ---------------------------------------------------------------------------------------------------( 2 ) 2 f 0 f HF ( L C ) R FB V P-P (EQ. 52) V P-P 2 f 0 f HF L C R FB R C = ----------------------------------------------------------------------------------------0.75 V IN ( 2 f HF L C - 1 ) 0.75 V IN ( 2 f HF L C - 1 ) C C = ---------------------------------------------------------------------------------------------------( 2 ) 2 f 0 f HF ( L C ) R FB V P-P
2
0.66 V IN C C = --------------------------------------------------2 V PP R FB f 0 1 1 ------------------------------- f 0 < -----------------------------------2 C ESR 2 LC V PP ( 2 ) 2 f 02 L C R C = R FB ---------------------------------------------------------------0.66 V
IN
Case 2:
(EQ. 51)
0.66 V IN C C = ------------------------------------------------------------------------------------( 2 ) 2 f 02 V PP R FB L C
Case 3:
1 f 0 > -----------------------------------2 C ESR 2 f 0 V P-P L R C = R FB --------------------------------------------0.66 V IN ESR 0.66 V IN ESR C C C = ---------------------------------------------------------------2 V PP R FB f 0 L
In the solutions to the compensation equations, there is a single degree of freedom. For the solutions presented in Equation 53, RFB is selected arbitrarily. The remaining compensation components are then selected according to Equation 53. In Equation 53, L is the per-channel filter inductance divided by the number of active channels; C is the sum total of all output capacitors; ESR is the equivalent-series resistance of the bulk output-filter capacitance; and VPP is the peak-topeak sawtooth signal amplitude as described in Electrical Specifications on page 6. Case 1:
1 ------------------------------- > f 0 2 LC 2 f 0 V P-P L C R C = R FB ---------------------------------------------------------0.66 V
IN
Compensation Without Loadline Regulation
The non load-line regulated converter is accurately modeled as a voltage-mode regulator with two poles at the L-C resonant frequency and a zero at the ESR frequency. A type III controller, as shown in Figure 23, provides the necessary compensation.
C2
RC
CC
COMP
0.66 V IN C C = --------------------------------------------------2 V PP R FB f 0 1 1 ------------------------------- f 0 < -----------------------------------2 C ESR 2 LC V PP ( 2 ) 2 f 02 L C R C = R FB ---------------------------------------------------------------0.66 V
IN
FB C1 R1 RFB VSEN
ISL6323A
Case 2:
(EQ. 53)
FIGURE 23. COMPENSATION CIRCUIT WITHOUT LOAD-LINE REGULATION
0.66 V IN C C = -------------------------------------------------------------------------------------2 f 2 V (2 ) 0 P-P R FB L C
The first step is to choose the desired bandwidth, f0, of the compensated system. Choose a frequency high enough to assure adequate transient performance but not higher than 1/3 of the switching frequency. The type-III compensator has an extra high-frequency pole, fHF. This pole can be used for added noise rejection or to assure adequate attenuation at the error-amplifier high-order pole and zero frequencies. A good general rule is to choose fHF = 10f0, but it can be higher if desired. Choosing fHF to be lower than 10f0 can cause problems with too much phase shift below the system bandwidth, 30
Case 3:
1 f 0 > -----------------------------------2 C ESR 2 f 0 V P-P L R C = R FB --------------------------------------------0.66 V IN ESR 0.66 V IN ESR C C C = ----------------------------------------------------------------2 V P-P R FB f 0 L
FN6878.0
ISL6323A
Output Filter Design
The output inductors and the output capacitor bank together to form a low-pass filter responsible for smoothing the pulsating voltage at the phase nodes. The output filter also must provide the transient energy until the regulator can respond. Because it has a low bandwidth compared to the switching frequency, the output filter limits the system transient response. The output capacitors must supply or sink load current while the current in the output inductors increases or decreases to meet the demand. In high-speed converters, the output capacitor bank is usually the most costly (and often the largest) part of the circuit. Output filter design begins with minimizing the cost of this part of the circuit. The critical load parameters in choosing the output capacitors are the maximum size of the load step, I, the load-current slew rate, di/dt, and the maximum allowable output-voltage deviation under transient loading, VMAX. Capacitors are characterized according to their capacitance, ESR, and ESL (equivalent series inductance). At the beginning of the load transient, the output capacitors supply all of the transient current. The output voltage will initially deviate by an amount approximated by the voltage drop across the ESL. As the load current increases, the voltage drop across the ESR increases linearly until the load current reaches its final value. The capacitors selected must have sufficiently low ESL and ESR so that the total output-voltage deviation is less than the allowable maximum. Neglecting the contribution of inductor current and regulator response, the output voltage initially deviates by an amount as shown in Equation 54:
di V ESL ---- + ESR I dt (EQ. 54)
.
V - N V OUT V OUT IN L ESR ------------------------------------------------------------------f S V IN V P-P( MAX )
(EQ. 55)
Since the capacitors are supplying a decreasing portion of the load current while the regulator recovers from the transient, the capacitor voltage becomes slightly depleted. The output inductors must be capable of assuming the entire load current before the output voltage decreases more than VMAX. This places an upper limit on inductance. Equation 56 gives the upper limit on L for the cases when the trailing edge of the current transient causes a greater output-voltage deviation than the leading edge. Equation 57 addresses the leading edge. Normally, the trailing edge dictates the selection of L because duty cycles are usually less than 50%. Nevertheless, both inequalities should be evaluated, and L should be selected based on the lower of the two results. In each equation, L is the per-channel inductance, C is the total output capacitance, and N is the number of active channels.
2 N C VO L --------------------------------- V MAX - ( I ESR ) ( I ) 2 1.25 N C L ---------------------------- V MAX - ( I ESR ) V IN - V O ( I ) 2 (EQ. 56)
(EQ. 57)
Switching Frequency
There are a number of variables to consider when choosing the switching frequency, as there are considerable effects on the upper MOSFET loss calculation. These effects are outlined in "MOSFETs" on page 25, and they establish the upper limit for the switching frequency. The lower limit is established by the requirement for fast transient response and small output-voltage ripple as outlined in "Output Filter Design" on page 31. Choose the lowest switching frequency that allows the regulator to meet the transient-response requirements. Switching frequency is determined by the selection of the frequency-setting resistor, RT. Figure 24 and Equation 58 are provided to assist in selecting the correct value for RT.
R T = 10 [10.61 - ( 1.035 log ( f S ) ) ] (EQ. 58)
The filter capacitor must have sufficiently low ESL and ESR so that V < VMAX. Most capacitor solutions rely on a mixture of high frequency capacitors with relatively low capacitance in combination with bulk capacitors having high capacitance but limited high-frequency performance. Minimizing the ESL of the high-frequency capacitors allows them to support the output voltage as the current increases. Minimizing the ESR of the bulk capacitors allows them to supply the increased current with less output voltage deviation. The ESR of the bulk capacitors also creates the majority of the output-voltage ripple. As the bulk capacitors sink and source the inductor AC ripple current (see "Interleaving" on page 11 and Equation 3), a voltage develops across the bulk capacitor ESR equal to IC(P-P)(ESR). Thus, once the output capacitors are selected, the maximum allowable ripple voltage, VP-P(MAX), determines the lower limit on the inductance.
31
FN6878.0
ISL6323A
1k
phase and two-phase designs respectively. Use the same approach for selecting the bulk capacitor type and number.
0.3 INPUT-CAPACITOR CURRENT (IRMS/IO)
IL(P-P) = 0 IL(P-P) = 0.25 IO
IL(P-P) = 0.5 IO IL(P-P) = 0.75 IO
RT (k)
100
0.2
0.1
10 10k
100k
1M
10M
SWITCHING FREQUENCY (Hz)
FIGURE 24. RT vs SWITCHING FREQUENCY
Input Capacitor Selection
The input capacitors are responsible for sourcing the AC component of the input current flowing into the upper MOSFETs. Their RMS current capacity must be sufficient to handle the AC component of the current drawn by the upper MOSFETs which is related to duty cycle and the number of active phases.
0.3 INPUT-CAPACITOR CURRENT (IRMS/IO) IL(P-P) = 0 IL(P-P) = 0.25 IO IL(P-P) = 0.5 IO IL(P-P) = 0.75 IO
0
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VIN/VO)
FIGURE 26. NORMALIZED INPUT-CAPACITOR RMS CURRENT FOR 3-PHASE CONVERTER
Low capacitance, high-frequency ceramic capacitors are needed in addition to the input bulk capacitors to suppress leading and falling edge voltage spikes. The spikes result from the high current slew rate produced by the upper MOSFET turn on and off. Select low ESL ceramic capacitors and place one as close as possible to each upper MOSFET drain to minimize board parasitics and maximize suppression.
0.3 INPUT-CAPACITOR CURRENT (IRMS/IO)
0.2
0.1
0.2
0
0.1
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VO/VIN)
IL(P-P) = 0 IL(P-P) = 0.5 IO IL(P-P) = 0.75 IO
0 0 0.2 0.4 0.6 0.8 1.0
FIGURE 25. NORMALIZED INPUT-CAPACITOR RMS CURRENT vs DUTY CYCLE FOR 4-PHASE CONVERTER
For a four-phase design, use Figure 25 to determine the input-capacitor RMS current requirement set by the duty cycle, maximum sustained output current (IO), and the ratio of the peak-to-peak inductor current (IL,PP) to IO. Select a bulk capacitor with a ripple current rating which will minimize the total number of input capacitors required to support the RMS current calculated. The voltage rating of the capacitors should also be at least 1.25 times greater than the maximum input voltage. Figures 26 and 27 provide the same input RMS current information for three-
DUTY CYCLE (VIN/VO)
FIGURE 27. NORMALIZED INPUT-CAPACITOR RMS CURRENT FOR 2-PHASE CONVERTER
Layout Considerations
MOSFETs switch very fast and efficiently. The speed with which the current transitions from one device to another causes voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade efficiency, radiate noise into the circuit and lead to device overvoltage stress. Careful component selection, layout, and placement minimizes these voltage spikes. Consider, as an example, the turnoff transition of the upper PWM MOSFET. Prior to turnoff, the upper MOSFET
FN6878.0
32
ISL6323A
was carrying channel current. During the turnoff, current stops flowing in the upper MOSFET and is picked up by the lower MOSFET. Any inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide circuit traces minimize the magnitude of voltage spikes. There are two sets of critical components in a DC/DC converter using a ISL6323A controller. The power components are the most critical because they switch large amounts of energy. Next are small signal components that connect to sensitive nodes or supply critical bypassing current and signal coupling. The power components should be placed first, which include the MOSFETs, input and output capacitors, and the inductors. It is important to have a symmetrical layout for each power train, preferably with the controller located equidistant from each. Symmetrical layout allows heat to be dissipated equally across all power trains. Equidistant placement of the controller to the CORE and NB power trains it controls through the integrated drivers helps keep the gate drive traces equally short, resulting in equal trace impedances and similar drive capability of all sets of MOSFETs. When placing the MOSFETs try to keep the source of the upper FETs and the drain of the lower FETs as close as thermally possible. Input high-frequency capacitors, CHF, should be placed close to the drain of the upper FETs and the source of the lower FETs. Input bulk capacitors, CBULK, case size typically limits following the same rule as the high-frequency input capacitors. Place the input bulk capacitors as close to the drain of the upper FETs as possible and minimize the distance to the source of the lower FETs. Locate the output inductors and output capacitors between the MOSFETs and the load. The high-frequency output decoupling capacitors (ceramic) should be placed as close as practicable to the decoupling target, making use of the shortest connection paths to any internal planes, such as vias to GND next or on the capacitor solder pad. The critical small components include the bypass capacitors (CFILTER) for VCC and PVCC, and many of the components surrounding the controller including the feedback network and current sense components. Locate the VCC/PVCC bypass capacitors as close to the ISL6323A as possible. It is especially important to locate the components associated with the feedback circuit close to their respective controller pins, since they belong to a high-impedance circuit loop, sensitive to EMI pick-up. A multi-layer printed circuit board is recommended. Figure 28 shows the connections of the critical components for the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer, usually the one underneath the component side of the board, for a ground plane and make all critical component 33 ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminal to output inductors short. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring.
Routing UGATE, LGATE, and PHASE Traces
Great attention should be paid to routing the UGATE, LGATE, and PHASE traces since they drive the power train MOSFETs using short, high current pulses. It is important to size them as large and as short as possible to reduce their overall impedance and inductance. They should be sized to carry at least one ampere of current (0.02" to 0.05"). Going between layers with vias should also be avoided, but if so, use two vias for interconnection when possible. Extra care should be given to the LGATE traces in particular since keeping their impedance and inductance low helps to significantly reduce the possibility of shoot-through. It is also important to route each channels UGATE and PHASE traces in as close proximity as possible to reduce their inductances.
Current Sense Component Placement and Trace Routing
One of the most critical aspects of the ISL6323A regulator layout is the placement of the inductor DCR current sense components and traces. The R-C current sense components must be placed as close to their respective ISEN+ and ISEN- pins on the ISL6323A as possible. The sense traces that connect the R-C sense components to each side of the output inductors should be routed on the bottom of the board, away from the noisy switching components located on the top of the board. These traces should be routed side by side, and they should be very thin traces. It's important to route these traces as far away from any other noisy traces or planes as possible. These traces should pick up as little noise as possible.
Thermal Management
For maximum thermal performance in high current, high switching frequency applications, connecting the thermal GND pad of the ISL6323A to the ground plane with multiple vias is recommended. This heat spreading allows the part to achieve its full thermal potential. It is also recommended that the controller be placed in a direct path of airflow if possible to help thermally manage the part.
FN6878.0
ISL6323A
RFB C2 CC RC FB COMP ISEN3+ ISEN3PWM3 RAPA CAPA APA DVC VSEN BOOT1 UGATE1 PHASE1 R1_1 LGATE1 R1_2 ISEN1ISEN1+ +12V +5V VCC CFILTER ROFS FS RFS RSET OFS BOOT2 UGATE2 PHASE2 R2_1 LGATE2 R2_2 R4_2 C2 CPU LOAD C4 V_CORE PVCC1_2 CFILTER CBOOT CBULK C HF +12V CIN CIN CBOOT BOOT2 VCC PVCC CFILTER UGATE2 GND PHASE2 R4_1 PWM2 LGATE2 ISL6614 C1 LGATE1 PWM1 PGND CBOOT +12V CIN R3_2 C3 R3_1CIN +12V CBOOT BOOT1 UGATE1 PHASE1
+12V
+5V RSET
NC NC
VFIXEN ISEN2SEL ISEN2+ SVD SVC RGND VID4 VID5 PWROK VDDPWRGD ISEN4+ GND ISEN4PWM4
+12V REN1 OFF ON REN2 EN
ISL6323A PVCC_NB
+12V
KEY HEAVY TRACE ON CIRCUIT PLANE LAYER CIN ISLAND ON POWER PLANE LAYER ISLAND ON CIRCUIT PLANE LAYER VIA CONNECTION TO GROUND PLANE
CFILTER CBOOT_NB
BOOT_NB UGATE_NB PHASE_NB
V_NB R1_NB C 1_NB
CHF CBULK
LGATE_NB ISEN_NBISEN_NB+ COMP_NB FB_NB C2_NB RC_NB CC_NB RFB_NB
R2_NB
RED COMPONENTS: LOCATE CLOSE TO IC TO MINIMIZE CONNECTION PATH BLUE COMPONENTS: LOCATE NEAR LOAD (MINIMIZE CONNECTION PATH) MAGENTA COMPONENTS: LOCATE CLOSE TO SWITCHING TRANSISTORS (MINIMIZE CONNECTION PATH)
NB LOAD
FIGURE 28. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation's quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com 34
FN6878.0
ISL6323A
Package Outline Drawing
L48.7x7
48 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 4, 10/06
4X 5.5 7.00 A B 6 PIN 1 INDEX AREA 37 36 44X 0.50 48 1 6 PIN #1 INDEX AREA
7.00
4. 30 0 . 15
25 (4X) 0.15 24 TOP VIEW 48X 0 . 40 0 . 1 13
12
0.10 M C A B 4 0.23 +0.07 / -0.05
BOTTOM VIEW
SEE DETAIL "X" 0.10 C BASE PLANE C
( 6 . 80 TYP ) ( 4 . 30 )
0 . 90 0 . 1
SIDE VIEW ( 44X 0 . 5 )
SEATING PLANE 0.08 C
C ( 48X 0 . 23 ) ( 48X 0 . 60 ) TYPICAL RECOMMENDED LAND PATTERN
0 . 2 REF
5
0 . 00 MIN. 0 . 05 MAX. DETAIL "X"
NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 indentifier may be either a mold or mark feature.
35
FN6878.0

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