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HV9925
Initial Release
Programmable-Current LED Lamp Driver IC with PWM Dimming
Features
Programmable Output Current to 50mA PWM Dimming / Enable Universal 85-264VAC Operation Fixed OFF-Time Buck Converter Internal 500V Power MOSFET Over Temperature Protection with Hysteresis
General Description
The HV9925 is a pulse width modulated (PWM) high-efficiency LED driver control IC with PWM dimming capabilities. It allows efficient operation of high brightness LED strings from voltage sources ranging up to 400VDC. The HV9925 includes an internal high-voltage switching MOSFET controlled with a fixed off-time TOFF of approximately 10s. The LED string is driven at constant current, thus providing constant light output and enhanced reliability. Selecting a value of a current sense resistor can externally program the output LED current of the HV9925. The peak current control scheme provides good regulation of the output current throughout the universal AC line voltage range of 85 to 264VAC or DC input voltage of 20 to 400V. The HV9925 is designed with a built in thermal shutdown to prevent excessive power dissipation in the IC.
Applications
Decorative Lighting Low Power Lighting Fixtures
Typical Application Circuit
~ AC ~ BR1 L1 CIN CO LEDN D1 LED1
6 Enable 3 4 CDD
Drain PWMD
7
Drain
8
Drain
HV9925
VDD RSENSE
U1
GND
1 RSENSE
2
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HV9925
Ordering Information
DEVICE HV9925
-G indicates package is RoHS compliant (`Green')
Package Options 8-Pin SOIC w/ Heat Slug HV9925SG-G
Absolute Maximum Ratings*
Parameter Supply Voltage, VDD Supply Current, IDD PWMD, RSENSE Voltage Operating Ambient Temperature Range Operating Junction Temperature Range Storage Temperature Range Power Dissipation @ 25C
All voltages referenced to GND pin. **The power dissipation is given for the standard minimum pad without a heat slug, and based on RJA=125C/W. RJA is the sum of the junction-tocase and case-to-ambient thermal resistance, where the latter is determined by the user's board design. The junction-to-ambient thermal resistance is RJA= 105C/W when the part is mounted on a 0.04 in2 pad of 1 oz copper, and RJA= 60C/W when mounted on a 1 in2 pad of 1 oz copper.
Pin Configuration
Value -0.3 to +10V -0.3 to +10V +5mA -40C to +85C -40C to +125C -65C to +150C 800mW**
RSENSE GND PWMD VDD
1 2 3 4
HV9925SG
8 7 6 5
Drain Drain Drain NC
top view SO-8 + Heat Slug
(Heat Slug Potential is at ground)
DRAIN (6,7,8) - This is a drain terminal of the output switching MOSFET and a linear regulator input. VDD (4) - This is a power supply pin for internal control circuits. Bypass this pin with a 0.1uF low impedance capacitor. RSENSE (1) - This is a source terminal of the output switching MOSFET provided for current sense resistor connection. GND (2) - This is a common connection for all circuits. PWMD (3) - This is the PWM Dimming input to the IC.
Electrical Characteristics
Symbol VDD VUVLO VUVLO IDD VBR VDRAIN RON CDRAIN ISAT VTH TBLANK TON(MIN) Parameter VDD regulator output
(The * denotes the specifications which apply over the full operating temperature range of -40C < TA < +85C, otherwise the specifications are at TA = 25C. VDRAIN = 100V, unless otherwise noted)
Min 5.0 500 20 100 0.44 200 -
Typ 7.5 200 300 100 1.0 150 0.47 300 -
Max 500 200 5.0 0.50 400 650
Units V V mV A V V pF mA V ns ns
Conditions ---VDD(EXT) = 8.5V * --IDRAIN = 50mA VDRAIN = 400V --* ---
VDD undervoltage threshold VDD undervoltage lockout hysteresis Operating supply current Breakdown voltage VDRAIN supply voltage ON resistance Output capacitance DRAIN saturation current Threshold voltage Leading edge blanking delay Minimum ON time
Output (DRAIN)
Current Sense Comparator
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Electrical Characteristics (cont.)
Symbol TOFF VPWMD,HI VPWMD,LO RPWMD TOT THYST Parameter OFF time PWMD input high voltage PWMD input low voltage PWMD pull down resistance Over temperature trip limit Temperature hysteresis Min 8.0 2.0 100 Typ 10.5 200 140 60 Max 13 0.8 300 Units s V V k C C Conditions -* -* -VPWMD = 5V --OFF-Time Generator PWM Dimming
Thermal Shutdown
Functional Block Diagram
GND VDD DRAIN
PWMD
REG
TOFF = 10s
0.5V
S + R
Q Q Over Temp.
TBLANK = 300ns
HV9925
RSENSE
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Typical Performance Characteristics (T
0.485
J
= 25OC unless otherwise noted)
200 180
Threshold Voltage VTH vs Temperature TJ
ON Resistance RON vs Temperature TJ
Current Sense Threshold, V
ON Resistance, Ohm
-40 -15 10 35 60 85 110
0.480
160 140 120 100 80 60
0.475
0.470
0.465
0.460
40 -40 -15
Junction Temperature, C
Junction Temperature, C
10
35
60
85
110
13.0 12.5
OFF Time TOFF vs Temperature TJ
1000
Output Capacitance CDRAIN vs VDRAIN
OFF Time, s
12.0 11.5 11.0 10.5 10.0 9.5 9.0 -40 -15
DRAIN Capacitance, pF
100
10
1
Junction Temperature, C
10
35
60
85
110
0
10
20
30
40
DRAIN Voltage, V Output Characteristics IDRAIN vs VDRAIN
580
DRAIN Breakdown Voltage BV vs TJ
180 160
DRAIN Breakdown Voltage, V
570
DRAIN Current, mA
560 550 540 530 520 510 500 490 -40 -15 10 35 60 85 110
140 120 100 80 60 40 20 0 0
TJ = 25OC TJ = 125OC
10
20
30
40
Junction Tem perature, C
DRAIN Voltage, V
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Functional Description
The HV9925 is a PWM peak current control IC for driving a buck converter topology in continuous conduction mode (CCM). The HV9925 controls the output current (rather than output voltage) of the converter that can be programmed by a single external resistor (RSENSE), for the purpose of driving a string of light emitting diodes (LED). An external enable input (PWMD) is provided that can be utilized for PWM dimming of an LED string. The typical rising and falling edge transitions of the LED current when using the PWM dimming feature of the HV9925 are shown in Fig. 6 and Fig. 7. When the input voltage of 20 to 400V appears at the DRAIN pin, the internal linear regulator seeks to maintain a voltage of 7.5VDC at the VDD pin. Until this voltage exceeds the internally programmed under-voltage threshold, no output switching occurs. When the threshold is exceeded, the integrated high-voltage switch turns on, pulling the DRAIN low. A 200mV hysteresis is incorporated with the undervoltage comparator to prevent oscillation. When the voltage at RSENSE exceeds 0.47V, the switch turns off and the DRAIN output becomes high impedance. At the same time, a one-shot circuit is activated that determines the off-time of the switch (10s typ.). A "blanking" delay of 300ns is provided upon the turn-on of the switch that prevents false triggering of the current sense comparator due to the leading edge spike caused by circuit parasitics. variation. Therefore, the output current will remain unaffected by the varying input voltage. Adding a filter capacitor across the LED string can reduce the output current ripple even further, thus permitting a reduced value of L1. However, one must keep in mind that the peak-to-average current error is affected by the variation of TOFF. Therefore, the initial output current accuracy might be sacrificed at large ripple current in L1. Another important aspect of designing an LED driver with HV9925 is related to certain parasitic elements of the circuit, including distributed coil capacitance of L1, junction capacitance, and reverse recovery of the rectifier diode D1, capacitance of the printed circuit board traces CPCB and output capacitance CDRAIN of the controller itself. These parasitic elements affect the efficiency of the switching converter and could potentially cause false triggering of the current sense comparator if not properly managed. Minimizing these parasitics is essential for efficient and reliable operation of HV9925. Coil capacitance of inductors is typically provided in the manufacturer's data books either directly or in terms of the self-resonant frequency (SRF). SRF = 1/(2 L CL ) where L is the inductance value, and CL is the coil capacitance. Charging and discharging this capacitance every switching cycle causes high-current spikes in the LED string. Therefore, connecting a small capacitor CO (~10nF) is recommended to bypass these spikes. Using an ultra-fast rectifier diode for D1 is recommended to achieve high efficiency and reduce the risk of false triggering of the current sense comparator. Using diodes with shorter reverse recovery time trr, and lower junction capacitance CJ, achieves better performance. The reverse voltage rating VR of the diode must be greater than the maximum input voltage of the LED lamp. The total parasitic capacitance present at the DRAIN output of the HV9925 can be calculated as: CP = CDRAIN + CPCB + CL + CJ (3) When the switch turns on, the capacitance CP is discharged into the DRAIN output of the IC. The discharge current is limited to about 150mA typically. However, it may become lower at increased junction temperature. The duration of the leading edge current spike can be estimated as: TSPIKE = VIN CP + trr ISAT (4)
Application Information
Selecting L1 and D1 The required value of L1 is inversely proportional to the ripple current IO in it. Setting the relative peak-to-peak ripple to 20~30% is a good practice to ensure noise immunity of the current sense comparator. V T L1 = O OFF (1) IO VO is the forward voltage of the LED string. TOFF is the offtime of the HV9925. The output current in the LED string (IO) is calculated then as: IO = VTH 1 - IO RSENSE 2 (2)
where VTH is the current sense comparator threshold, and RSENSE is the current sense resistor. The ripple current introduces a peak-to-average error in the output current setting that needs to be accounted for. Due to the constant off-time control technique used in the HV9925, the ripple current is nearly independent of the input AC or DC voltage
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HV9925
In order to avoid false triggering of the current sense comparator, CP must be minimized in accordance with the following expression: CP < ISAT TBLANK (MIN) - trr VIN(MAX ) rectified AC line input, the exact equation for calculating the conduction loss is more cumbersome. However, it can be estimated using the following equation: (10) P = K I 2 R + K I V
COND C O ON d DD AC
(
)
(5)
where TBLANK(MIN) is the minimum blanking time of 200ns, and VIN(MAX) is the maximum instantaneous input voltage. The typical DRAIN and RSENSE voltage waveforms are shown in Fig. 3 and Fig. 4. Estimating Power Loss Discharging the parasitic capacitance CP into the DRAIN output of the HV9925 is responsible for the bulk of the switching power loss. It can be estimated using the following equation: PSWITCH C V 2 = P IN + VINISAT trr FS 2 (6)
where VAC is the input AC line voltage. The coefficients KC and Kd can be determined from the minimum duty ratio Dm=0.71Vo/(VAC).
0.7
0.6
0.5 Kd( Dm) Kc( Dm) 0.4
0.3
where Fs is the switching frequency and ISAT is the saturated DRAIN current of the HV9925. The switching loss is the greatest at the maximum input voltage. The switching frequency is given by the following: FS = VIN - VO VIN TOFF
-1
0.2
0.1
0
0.1
0.2
0.3 Dm
0.4
0.5
0.6
0.7
(7)
Fig. 1. Conduction Loss Coefficients KC and Kd EMI Filter As with all off-line converters, selecting an input filter is critical to obtaining good EMI. A switching side capacitor, albeit of small value, is necessary in order to ensure low impedance to the high frequency switching currents of the converter. As a rule of thumb, this capacitor should be approximately 0.10.2 F/W of LED output power. A recommended input filter is shown in Figure 2 for the following design example. Design Example 1 Let us design an HV9925 LED lamp driver meeting the following specifications: Input: Universal AC, 85-264VAC Output Current: 20mA Load: String of 10 LED (LW541C by OSRAM VF = 4.1V max. each) The schematic diagram of the LED driver is shown in Fig.2.
where is the efficiency of the power converter. When the HV9925 LED driver is powered from the full-wave rectified AC input, the switching power loss can be estimated as: (8) 1 PSWITCH ( VAC CP + 2 ISAT trr ) VAC - -1 VO 2 TOFF
(
)
VAC is the input AC line voltage. The switching power loss associated with turn-off transitions of the DRAIN output can be disregarded. Due to the large amount of parasitic capacitance connected to this switching node, the turn-off transition occurs essentially at zerovoltage. Conduction power loss in the HV9925 can be calculated as: (9) PCOND = D IO2 RON + IDD VIN (1 - D ) where D = VO /VIN is the duty ratio, RON is the ON resistance, IDD is the internal linear regulator current. When the LED driver is powered from the full-wave
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HV9925
Step 1. Calculating L1. The output voltage VO = 10 * VF 41V (max.). Use equation (1) assuming a 30% peak-to-peak ripple. L1 = 41V 10s = 68mH 0.3 20mA Step 6. Selecting input capacitor CIN Output Power = 41V 20mA = 820mW Select CIN ECQ-E4104KF by Panasonic (0.1F, 400V, Metalized Polyester Film). Design Example 2 Select L1 68mH, I=30mA. Typical SRF = 170KHz. Calculate the coil capacitance.
CL = 1 L1 (2 SRF)
2
=
1 68mH (2 170KHz )2
13pF
Let us now design a PWM-dimmable LED lamp driver using the HV9925: Input: Universal AC, 85-135VAC Output Current: 50mA Load: String of 12 LED (Power TOPLED(R) by OSRAM, VF = 2.5V max. each) The schematic diagram of the LED driver is shown in Fig.3. We will use an aluminum electrolytic capacitor for CIN in order to prevent interruptions of the LED current at zero crossings of the input voltage. As a"rule of thumb", 2~3F per each watt of the input power is required for CIN in this case. Step 1. Calculating L1. The output voltage VO = 12 * VF = 30V (max.). Use equation (1) assuming a 30% peak-to-peak ripple. 30 V 10.5s L1 = = 21mH 0.3 50mA Select L1 22mH, I = 60mA. Typical SRF = 270KHz. Calculate the coil capacitance. 1 1 CL = = 15pF 2 22mH (2 270KHz )2 L1 (2 SRF) Step 2. Selecting D1 Select D1 ES1G with VR = 400V, trr 35ns and CJ < 10pF. Step 3. Calculating total parasitic capacitance using: (3) CP = 5pF + 5pF + 13pF + 8pF = 31pF Step 4. Calculating the leading edge spike duration using (4), (5) TSPIKE = 135 V 2 35pF + 35ns 100ns < TBLANK (MIN) 100mA
Step 2. Selecting D1 Usually, the reverse recovery characteristics of ultrafast rectifiers at IF = 20~50mA are not provided in the manufacturer's data books. The designer may want to experiment with different diodes to achieve the best result. Select D1 MUR160 with VR = 600V, trr 20ns (IF = 20mA, IRR = 100mA) and CJ 8pF (VF>50V). Step 3. Calculating total parasitic capacitance using: CP = 5pF + 5pF + 13pF + 8pF = 31pF (3)
Step 4. Calculating the leading edge spike duration using: (4), (5)
TSPIKE = 264 V 2 31pF + 20ns 136ns < TBLANK (MIN) 100mA
Step 5. Estimating power dissipation in HV9925 at 264VAC using (8) and (10) Let us assume that the overall efficiency = 0.7. Switching power loss:
PSWITCH 41V 1 ( 264V 31pF + 2 100mA 20ns ) 264V - 0 .7 2 10s
PSWITCH 125mW Minimum duty ratio: Dm = 0.71 41V /(0.7 264 V ) 0.16 Conduction power loss:
PCOND = 0.25 ( 20mA ) 210 + 0.63 200A 264 V 55mW
2
Step 5. Estimating power dissipation in HV9925 at 135VAC using (6), (7) and (9) Switching power loss: FS = 135 V 2 - 30 V / 0.7 135 V 2 10s = 78kHz
Total power dissipation at VAC(max): PTOTAL = 125mW + 55mW = 180mW
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HV9925
PSWITCH = 35pF (135 V ) + 135 V 2 100mA 35ns 78kHz
2
(
)
Total power dissipation in HV9925: PTOTAL = 52mW + 217mW = 269mW Step 6. Selecting input capacitor CIN Output Power = 30 V 50mA = 1.5 W Select CIN 3.3F, 250V.
PSWITCH 52mW Minimum duty ratio: Dm = 30 V /(0.7 135 V 2 ) 0.23 Conduction power loss: PCOND = 30 V (50mA )2 200 0.7 85 V 2 30 V + 0.5mA 85 V 2 - 0 .7
PCOND = 217mW
Figure 2. Universal 85-264VAC LED Lamp Driver
(IO = 20mA, VO = 50V) from Example 1
D2 D4
D3 CIN2 D5
LIN CIN CO D1
LED1 -LED10
AC Line 85-264V
VRD1 F1
3
8
HV9925
U1
1 2
7 6
L1
4
CDD
RSENSE
Figure 3. 85-135VAC LED Lamp Driver with PWM Dimming
D2 D4 D3 D5 LED1 -LED12 CIN CO D1 R1
3 100~200Hz 4
AC Line 85-135V
L1
HV9925
U1
1 2
8 7 6
CDD
RSENSE
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Figure 4. Switching Waveforms. CH1: VRSENSE, CH2: VDRAIN Figure 5. Switch-On Transition - Leading Edge Spike. CH1: VRSENSE, CH2: VDRAIN
Figure 6. PWM Dimming - Rising Edge. CH4: 10xIOUT
Figure 7. PWM Dimming - Falling Edge. CH4: 10xIOUT
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HV9925
8-LEAD SMALL OUTLINE PACKAGE WITH HEAT SLUG (SG)
0.1935 +/- 0.0035 (4.915 +/- 0.085)
0.1 +/- 0.01 (2.54 +/- 0.25)
Heat Slug
0.1535 +/- 0.0035
0.236 +/- .008
(3.9) (5.995) (+/- 0.09) (+/- 0.205)
0.0165 +/- 0.0035 (0.42 +/- 0.09) 0.14 +/- 0.01 (3.555 +/- 0.255)
0.055 +/- 0.005 (1.395 +/- 0.125)
0.0085 +/- 0.0015 (0.215 +/- 0.035)
0.05 +/- 0.01 (1.27 +/- 0.25)
0.0575 +/- 0.0065 (1.46 +/- 0.16)
0.0015 +/- .0025 (0.065 +/- 0.035)
0.033 +/- 0.017 (0.84 +/- 0.43)
Measurement Legend =
Dimensions in Inches (Dimensions in Millimeters)
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